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Analysis, design and implementation of a high step‐up multi‐port non‐isolated converter with coupled inductor and soft switching for photovoltaic applications

Analysis, design and implementation of a high step‐up multi‐port non‐isolated converter with... INTRODUCTIONOne of the most important factors in the economic growth and development of a country in the modern world is the proper use and optimal consumption of energy [1–5]. In recent decades, the use of renewable energy has expanded significantly due to the declining and depletion of fossil fuels and concerns about environmental pollution [6, 7].One of the largest sources of energy for the earth is the sun, the energy of which is used in various ways [8, 9]. Solar power generation systems are mainly capable of generating energy during the day and incur high costs if energy storage is required for future use [10, 11]. Another type of system that feeds from renewable sources is wind turbines. One of the disadvantages of these systems is the need for installation in geographical windbreaks [12, 13]. Other energy generation systems include fuel cells. Fuel cells are a new technology for generating energy that produces high‐efficiency electrical energy from a direct combination of fuel and oxidizer without causing environmental and noise pollution. In fuel cells, hydrogen gas is mainly used as fuel and from its reaction with oxygen, in addition to electrical energy, water and heat are also produced. The main problems of fuel cells are the slow dynamics of the fuel cell and large changes in output voltage with load changes [14]. At the moment, due to the high cost of the main element of these systems, the use of this type of energy production system alone is not economical. Usually, it is necessary to use dc‐dc converters to adapt to variable output voltage, renewable energy sources with load voltage (dc bus) and also to manage power in these sources [15, 16]. By considering the strengths and weaknesses of each of these systems, it can be concluded that by combining energy production systems, in addition to using the strengths of each, their weaknesses can be minimized [17, 18]. Due to the variety of energy sources and the need to use more than one power source in some applications, it is better to use a multi‐input converter instead of several independent converters because in multi‐input converters use less semiconductor elements, less inductors and fewer capacitors. Therefore, its control structure is much simpler compared to several single‐input converters, which has increased the efficiency and increased the energy conversion capacity, as well as reduced the cost of manufacturing the converter [19, 20]. Due to the high importance of overall system efficiency in renewable energy production systems and also the limited space in many applications, increasing the efficiency and reducing the volume of these converters is very desirable. In many applications of multi‐input converters, especially in solar systems where the output voltage of the array terminal should not be more than a certain value, non‐isolated incremental converters can be used to provide high voltage gain and high efficiency at the same time [21].Types of multi‐input convertersIn a general classification, the types of multi‐input converters can be classified into two types of isolated multi‐input converters [22, 23]. and non‐isolated multi‐input converters [24, 25]. According to Figure 1. Isolation in multi‐input converters has advantages and disadvantages that should be selected based on the application of the system. Its advantage is the isolation of the sources from each other (in multi‐input magnetic converters) as well as the load of the sources, which allows the use of different sources with different voltages. Disadvantages of isolation include design problems for transformers with multiple windings as well as increased circuit volume. According to the above, the presence or absence of isolation depends entirely on the design of the system and is not merely a unique criterion for the suitability or inappropriateness of the converter [26]. In the non‐isolated converter topology, all ports have a common ground. High step‐up non‐isolation multi‐input dc‐dc converters are used in a variety of applications. Features of this converter are low number of parts, flexibility to increase the number of input sources, provide energy of power storage system and increase voltage gain. These converters use soft switching methods to increase the efficiency of the converter. One of the most important factors to consider when designing multi‐input converters is reducing the number of converter components. Non‐isolated multi‐input converters are more attractive for certain applications compared to isolated multi‐input converters due to the simplicity of structure and high efficiency as well as lower cost in applications where high voltage gain and isolation are not required [27].1FIGUREIsolation in multi‐port convertersNon‐isolated multi‐input converters are obtained by serial or parallel connection of base converters [28, 29]. Refs. [30] and [31] provide a complete analysis of how to extract different types of non‐isolated multi‐input converter structures. In this analysis, the types of non‐isolated multi‐input converter structures are classified into two general categories: pulse voltage source cell‐based (PVSC) multi‐input converters and pulse current cell‐based (PCSC) multi‐input converters. This analysis is the basis for extracting a variety of non‐isolated multi‐input converter structures.In ref. [32] different types of multi‐input converter topology with the help of these synthesis processes using PVSC and (PCSC) basic converters based on pulse width modulation (PWM) are shown. It has also been suggested that the current pulse source with the help of a boost converter can be suitable for high gain dc‐dc converters for hybrid electric vehicles.In ref. [33] a general method of synthesizing multi‐input converters using base converter cells, in other words PCSC and output filter cells is introduced. In this reference, a multi‐input converter with PCSC connection with a sample of output filter cells is proposed for which it does not require any energy buffer step. As a result, this topology saves on construction costs as well as simplifies design. Also, to better understand the above concepts, two design examples are given to illustrate the synthesis of multi‐input converters for use in power supply applications, Internet‐of‐Things (IOT) devices, and other applications.It can be said one of the most important factors that could be assumed in design and implementation of the most multi‐input converters is reducing the number of components.Soft switching in multi‐input convertersUsing multi‐input converters, it will be possible to remove some common elements in two separate converters. As a result, the number of general structure elements will decrease. So it will be relatively easier to provide soft switching in the resulting structure compared to two separate converters [34, 35].Based on the method used to provide soft switching in non‐isolated multi‐input converters, these converters classified into three general categories: non‐isolated resonant multi‐input converters [36], non‐isolated multi‐input converters with active clamp [37]. and non‐isolated multi‐input converters with soft switching [38].In multi‐input resonant converters with proper converter design, by adding additional elements, current and voltage pulses are transmitted in sinusoidal or quasi‐sinusoidal form. Using these methods, soft switching conditions and effective reduction of electromagnetic interference are provided. One of the problems with this method is the high stress current and voltage of the circuit elements. Also, in order to provide soft switching conditions for a wide range of load and input changes, there is mainly a need to use variable control methods, which will complicate the control circuit. In the multi‐input converter with active clamp, the at zero voltage switching conditions (ZVS) are provided by adding only one auxiliary switch and a small number of passive auxiliary elements [39]. In these converters, contrary resonant converters, the voltage stress and current stress of the elements are within the minimum value range. Also, it is possible to control the converter with conventional methods of controlling pulse width modulation and with constant frequency. However, the main problem with this method is the high eddy current of the converter. This eddy current is almost constant for a wide range of load changes, which reduces efficiency, especially at low powers. In multi‐input converters whose inputs are mainly renewable sources and also have large changes in production power during the day, this factor causes weakness in these converters.In soft‐switching multi‐input converters, zero‐voltage switching conditions are provided for a wide range of output power variations with minimal circulation current. Due to the advantages and disadvantages of the proposed methods, soft switching multi‐input converters have many advantages. Also, the voltage stress of the elements is minimal and the converter can be controlled by conventional on pulse width modulation control methods for a wide range of power and voltage changes. However, the main problem with multi‐input soft‐switching converters is the large number of auxiliary elements of the converter [40].Most energy sources are inherently low voltage and high step‐up structures are used to increase their voltage gain. To increase the voltage, gain, different methods are used in different converters according to their type and application. Some of these structures include coupled inductors, series capacitors in the power flow path, isolated transformers, and switched diode‐capacitor structures. Using these methods solves the problems associated with the high duty cycle in a conventional boost converter and improves the performance of the converter. Leakage inductance of coupled inductors is used to control the diode current drop and also to reduce the problem of reverse diode recovery and some soft switching methods are implemented by the coupled inductor to reduce switching losses. also the coupled inductor can act as a transformer to increase the voltage in non‐isolated high step‐up converters to avoid a very high duty cycle [41]. The capacitor is used as another voltage source to achieve a high gain [42].In general, it can be mentioned that the use of high frequencies in circuits increases switching losses and reduces circuit efficiency. Therefore, in order to reduce losses and increase the frequency, the use of soft circuit switching techniques is unavoidable. On the other hand, using soft switching techniques is a good solution to reduce RFI and EMI noise.Literature surveyHigh step‐up non‐isolated dc‐dc converters have a variety of applications. One of the most important features of these converters is the reduction of converter elements, flexibility to increase the number of input sources, power supply of power storage system and increase voltage gain [43, 44]. These converters use soft switching methods to increase the efficiency of the converter. One of the most important factors to consider when designing multi‐input converters is reducing the number of converter components.A general method for obtaining high step‐up non‐isolated dc‐dc converters using conventional dc‐dc converters with a circuit consisting of an inductor and voltage multiplier cells is presented in ref. [45].A high step‐up dc‐dc converter is proposed in ref. [46], Based on a combined structure of the coupled inductor and the switched‐capacitor, which simple structure and so low voltage switches help to increase the efficiency of the proposed converter. In addition, the voltage stress of capacitors and diodes is significantly reduced. Lack of soft switching and so the lack of multipolar capability of this topology are the problems of this proposed structure.A high‐gain dc‐dc converter with soft switching, based on the coupled inductor and switched‐capacitor technique, is provided in ref. [47]. capacitors are used in this converter that are charged in parallel and discharged in series with a coupled inductor to achieve high voltage gain. Using the energy stored in the inductance leakage of the coupled inductor, both switches act as ZVS, which increases efficiency. One of the disadvantages of this converter is the inability of the proposed multipolar topology.A high step‐up dc‐dc converter has been provided in ref. [48] that combines an inductor structure with a voltage doubling circuit to increase the voltage gain to achieve a high voltage gain. This proposed converter can be used effectively for the structure of a modular dc‐dc converter for solar cell systems. The only problem with this converter is the inability of the multipolar topology.A dc‐dc converter with high step‐up three‐ports is provided in ref. [49], which provides two separate paths for transferring power from the input source to the output load, in fact, for each input, it has a separate phase, so that the components work in each mode of operation changes. also It has a port for energy storage device such as batteries and ultra‐capacitors that each input source and energy storage device can provide a unique path of power current to the output load directly from the power source regardless of the load power status. Therefore, in order to reduce the number of converter components, some components are shared in different operating modes. the technique of coupled inductors is used to increase the voltage in this converter and also, to reduce the effect of leakage inductance and to create soft switching conditions, two active clamp circuits have been used. This type of converter has the ability to connect to other high step‐up converters, which include the coupled inductor and the active clamp structure. it turns them into multi‐input converters, which are achieved while providing soft switching mode and eliminating of leakage inductor effect.A non‐isolated high step‐up dc‐dc converter is proposed to combine the coupled inductor structure with the voltage multiplier and also the switched‐capacitor structure to increase the voltage. Soft switching methods are used to increase the converter efficiency, and switched capacitor cells and the proposed converter voltage multiplier circuits can reduce the voltage stress of switches and diodes. This converter is designed for clean energy applications and can be used well in high efficiency applications. One of its limited disadvantages is the high volume of the converter and the large number of converter elements, which complicates the converter and another disadvantage is the lack of multipolar capability of this topology [50].A non‐isolated high gain converter is presented in ref. [51]. which increases the voltage gain of the converter by using a coupled inductor and two switched‐capacitors that are charged in parallel by the inductor and discharged in series also reduces losses in the circuit. The only problem with this converter is the inability of the multipolar topology.Actually, the use of high step‐up techniques solves the problems associated with the high duty cycle in a conventional incremental converters and improves the performance of the converters.Main study and innovationIn this paper, the structure of a non‐isolated multi‐input converter has proposed, which can be increased by using a combination of coupled inductor technique with switched‐capacitor. To reduce the leakage inductance effects and provide soft switching mode, two active clamp circuits have been used in this converter. Simulation results in different conditions have presented to evaluate and analyse the performance of the proposed multi‐input converter. Also, a laboratory sample of this proposed converter has been made to evaluate the results, which shows the comparison of the correct design and analysis results. In summary, the innovation of the article can be expressed as follows:Using two active clamp circuits to reduce leakage inductance effects.Simple structure and increase the converter voltage gain.Using a combination of two methods, coupled inductor and switched‐capacitor.Using switches with low voltage stress resulting in low conductivity.Reduction of voltage stress in the proposed converter.Making a laboratory sample to review the analysis and design of the converter.Main structure of this paperThe structure of the article is as follows. After stating the importance of the issue in the introduction, describes the second part of proposed non‐isolated high step‐up dc‐dc converter and its eight operating modes. In the third part, the converter design is mentioned in four sections, which the computation of static voltage gains and the computation of voltage and current stress have expressed. In the fourth part, the results of converter simulation using ORACD software for different modes have shown. In the fifth part, the laboratory results of the proposed converter in EES discharge and charge modes have given. In the sixth part, the conclusion of the article is given.PROPOSED CONVERTER STRUCTURE AND CONVERTER OPERATING MODESIn multi‐input converters, the type of isolation is selected based on the type of system application, which its advantages include, isolation of sources from each other (in multi‐input magnetic converters). as well as the source load that provides the use of different sources with different voltages. In general, the problems of designing transformers with multiple windings as well as increasing the circuit volume are the disadvantages of isolation. Due to the high voltage gain of the proposed converter and also due to the high efficiency and simple structure in this type of applications, non‐isolated converters have more used. The main block diagram of the high step‐up multi‐input converter is shown in Figure 2. The non‐isolated high step‐up dc‐dc proposed converter structure is shown in Figure 3. According to the structure of the converter, it can be seen that in order to increase the voltage gain in the proposed converter, the combination of the structure of the coupled inductor with the voltage multiplier has been used. Also, to reduce the voltage stress of the main switch due to leakage inductor energy, active clamp circuits have been used. By combining these two methods, switches with low voltage stress and consequently low conductivity can be used. Adding a voltage multiplier cell increases the amount of leakage inductor and the transformer turns ratio, and as a result reducing the circuit volume. The performance of the proposed converter depends on the charging or discharging energy storage system (ESS) mode. The following is a complete analysis of each of the ESS charging or discharging modes.2FIGUREHigh step‐up multi‐input converter block diagram3FIGUREProposed non‐isolated high step‐up multi‐input converter structureESS discharging modeIn this case, diode D is always off and both phases operate independently of each other. Energy is transferring from input to output and in fact both of the phases operate completely independently of each other but similar to each other. S1 and S2 switches act as the main switch of converter. Two active clamp circuits, including the S3 switch and the CC1 clamp capacitor in the upper phase, and so S4 switch and the CC2 clamp capacitor in the lower phase, are used to reduce the voltage stress of the main switch due to the effects of recover the leakage inductance and also provide soft switching conditions.Also, elements C1, C2, L2, Dd1 and Dd3 in the upper phase and elements C3, C4, L4, Dd2 and Dd4 in the lower phase are used to increase the voltage gain. C1, C2, C3 and C4 are snubber capacitors and L1, L2, L3 and L4 are coupled inductors. Because in this case, both phases operation is completely similar to each other, only the upper‐phase operation modes have been investigated and analysed.But due to the exact similarity of the upper and lower phases in the ESS discharge mode, the operating modes for the lower phase are exactly the same as the upper phase. Figure 4 shows the main key waveform diagrams in the proposed converter. According to Figure 5, the proposed converter in one switching period has eight operating modes, which are expressed in the following equation of each mode:4FIGUREMain key waveform diagrams in the proposed converter in ESS discharging mode5FIGUREOperating modes in ESS discharging mode of proposed converterMode 1—Time range (t0 – t1)By the switch is turned on, the output diode current continues to flow. In this time range, equations are expressed as follows:1ilm(t0)=ilm(t0)+2Vc1−Vonlm(t−t0)\begin{equation}{i_{lm}}({t_0}) = {i_{lm}}({t_{\rm{0}}}) + \frac{{2\,{V_{c{\rm{1}}}} - {V_o}}}{{n\,{l_m}}}(t - {t_{\rm{0}}})\end{equation}2ilk(t0)+Vin−Vlmllk(t−t0)=ilk(t0)+Vin−2Vc1−Vonllk(t−t0)\begin{equation}{i_{lk}}({t_0}) + \frac{{{V_{in}} - {V_{lm}}}}{{{l_{lk}}}}(t - {t_{\rm{0}}}) = {i_{lk}}({t_0}) + \frac{{{V_{in}} - \left(\frac{{2\,{V_{c1}} - {V_{\rm{o}}}}}{n}\right)}}{{{l_{lk}}}}(t - {t_{\rm{0}}})\end{equation}where Vc1 and Vc2 are the multiplier capacitors (voltage amplifier), respectively. In general, C1 and C2 and L2 and diodes Dd3 and Dd1 form the voltage multiplier circuit together. The leakage inductor current is ilk and the magnetic induction current is ilm, and the voltages Vin, Vo, and Vlm are the input, output, and double‐ended voltages of the magnetic inductor, respectively. Also n indicates the turns ratio and LLK is leakage inductance.Mode 2—Time range (t1 – t2)This mode starts the moment the output diode turns off while the S1 is still on, the iLK current reaches iLm and the iLK exceeds iLm. In this case, the following equation is established:3ilm(t)=ilm(t1)+Vc1nlm(t−t1)\begin{equation}{i_{lm}}(t) = {i_{lm}}({t_1}) + \frac{{\,{V_{c{\rm{1}}}}}}{{n\,{l_m}}}(t - {t_1})\end{equation}4VLK=Vin−Vlm\begin{equation}{V_{LK}} = {V_{in}} - {V_{lm}}\end{equation}5iLK(t)=iLK(t1)+nVin−Vc1nLLK(t−t1)\begin{equation}{i_{LK}}(t) = {i_{LK}}({t_1}) + \frac{{n{V_{in}} - {V_{c1}}}}{{n\,{L_{LK}}}}(t - {t_1})\end{equation}Mode 3—Time range (t2–t3)When this mode starts, the main switch turns off under zero voltage switching (ZVS). Leakage inductance begins to resonance with capacitors Cs1 and Cs3. As a result, VCs1 increases and VCs3 decreases. For this mode, the equations are expressed as follows:6ilm(t)=ilm(t2)+vc1(nlm)(t−t2)\begin{equation}{i_{lm}}(t) = {i_{lm}}({t_2}) + \frac{{{v_{c1}}}}{{(n\,{l_m})}}(t - {t_2})\end{equation}7iLK(t)=1ZV1−vc1n+1ZCs3Cs1+Cs3(Vcs3(t2)−Vcc)×sin(ω(t−t2)+iLK(t2)cosω(t−t2\begin{eqnarray} {i_{LK}}(t) &=& \left[\dfrac{1}{Z}\left({V_1} - \dfrac{{{v_{c1}}}}{n}\right) + \dfrac{1}{Z}\dfrac{{{C_{s3}}}}{{{C_{s1}} + {C_{s3}}}}({V_{cs3}}({t_2}) - {V_{cc}})\right]\nonumber\\ &&\times\, \sin (\omega (t - {t_2}) + {i_{LK}}({t_2})\,\cos \omega (t - {t_2}\end{eqnarray}8Vcs1(t)=V1−vc1n+vc1n−V1+CS3CS1+CS3(Vcc−Vcs3(t2))×cos(ωt2)+[ilk(t2)Z]sinω(t−t2)\begin{eqnarray} {V_{cs1}}(t) &=& {V_1} - \dfrac{{{v_{c1}}}}{n}\nonumber\\ && +\, \left[\left(\dfrac{{{v_{c1}}}}{n} - {V_1}\right) + \dfrac{{{C_{S3}}}}{{{C_{S1}} + {C_{S3}}}}({V_{cc}} - {V_{cs3}}({t_2}))\right]\nonumber\\ &&\times\, \cos (\omega {t_2}) + [{i_{lk}}({t_2})Z]\,\sin \omega (t - {t_2})\end{eqnarray}Resonance impedance (Zo) and angular resonance frequency (ωo) are equal to:9Z0=LLK(Cs1+Cs3)\begin{equation}{Z_0} = \sqrt {\frac{{{L_{LK}}}}{{({C_{s1}} + {C_{s3}})}}} \end{equation}10ω0=1(Cs1+Cs2)LLK\begin{equation}{\omega _0} = \frac{1}{{\sqrt {({C_{s1}} + {C_{s2}})\,{L_{LK}}} }}\end{equation}Mode 4—Time range (t3–t4)This mode is a continuation of the previous mode and VCs3 reaches zero. VCs1 reaches VCc1 and from now on only the diode corresponding to the S3 switch works. at the end of this interval, iLK decreases and reaches iLm and decreases. The equations for this mode of operation are as follows:11ilm(t)=ilm(t3)+vc1nlm(t−t3)\begin{equation}{i_{lm}}(t) = {i_{lm}}({t_3}) + \frac{{{v_{c1}}}}{{n{l_m}}}(t - {t_3})\end{equation}12ilk(t)=vi−vc1n−VccLlk(t−t3)+ilk(t3)\begin{equation}{i_{lk}}(t) = \left(\frac{{{v_i} - \frac{{{v_{c1}}}}{n} - {V_{cc}}}}{{{L_{lk}}}}\right)(t - {t_3}) + {i_{lk}}({t_3})\end{equation}Mode 5—Time range (t4 – t5)Because the amount of iLK current is less than iLm in this mode, the direction of the current changes (the polarity of the coupled inductor changes). By changing the polarity, Dd1 and Dd3 are turned off and the output diode Do1 is biased directly. the equations are established in this mode below:13ilm(t)=−Vo+2Vc1+VccnLm(t−t4)+ilm(t4)\begin{equation}{i_{lm}}(t) = \frac{{ - {V_o} + 2\,{V_{c1}} + {V_{cc}}}}{{n\,{L_m}}}(t - {t_4}) + {i_{lm}}({t_4})\end{equation}14ilk(t)=−Vcc−−Vo+2Vc1+Vccn+V1Llk(t−t4)+ilk(t4)\begin{equation}{i_{lk}}(t) = \frac{{ - {V_{cc}} - \left(\frac{{ - {V_o} + 2\,{V_{c1}} + {V_{cc}}}}{n}\right) + \,{V_1}}}{{{L_{lk}}}}(t - {t_4})+ {i_{lk}}({t_4})\end{equation}Mode 6—Time range (t5 – t6)During this interval, while the body diode of S3 is operating, auxiliary switch S3 turns on under ZVS conditions. current will pass through the diode, as long as the switch current is positive and as soon as it becomes negative, current will pass through the switch. in this mode the equations are established below:15ilm(t)=−Vo+2vc1+vccnLm)(t−t5)+ilm(t5)\begin{equation}{i_{lm}}(t) = \frac{{ - {V_o} + 2\,{v_{c1}} + {v_{cc}}}}{{n\,{L_m}}})(t - {t_5}) + {i_{lm}}({t_5})\end{equation}16ilk(t)=−Vcc−−Vo+2vc1+vccn+V1Llk)(t−t5)+ilk(t5)\begin{equation}{i_{lk}}(t) = \frac{{ - {V_{cc}} - \left(\frac{{ - {V_o} + 2\,{v_{c1}} + {v_{cc}}}}{n}\right) + {V_1}}}{{{L_{lk}}}})(t - {t_5})+ {i_{lk}}({t_5})\end{equation}Mode 7—Time range (t6–t7)This mode starts when the auxiliary switch S3 is turned off under ZVS conditions. Capacitors Cs1 and Cs3 are discharged and the value of VCs3 increases and VCs1 decreases. the capacitor voltage of the main switch VCs1 reaches zero at the end of the end of this mode and the body diode begins to conduct. In this mode, equations are expressed as below:17ilm(t)=−Vo+2Vc1+VccnLm)(t−t6)+ilm(t6)\begin{equation}{i_{lm}}(t) = \frac{{ - {V_o} + 2\,{V_{c1}} + {V_{cc}}}}{{n\,{L_m}}})\,(t - {t_6}) + {i_{lm}}({t_6})\end{equation}18ilk(t)=−ilm(t6)n+1)+2Vc1−Vo+nV1Z(n+1)+Cs1Cs1+Cs3Vcs3(t6)Z+Cs3Cs1+Cs3VccZsinω(t−t6)+ilm(t6)n+1+ilk(t6)cosω(t−t6)\begin{eqnarray} {i_{lk}}(t) &=& - \dfrac{{{i_{lm}}({t_6})}}{{n + 1}}) + \left[\dfrac{{2\,{V_{c1}} - {V_o} + n\,{V_1}}}{{Z\,(n + 1)}}\,+ \dfrac{{{C_{s1}}}}{{{C_{s1}} + {C_{s3}}}}\dfrac{{{V_{cs3}}({t_6})}}{Z}\right.\nonumber\\ &&+\, \left.\dfrac{{{C_{s3}}}} {{{C_{s1}} + {C_{s3}}}} \dfrac{{{V_{cc}}}}{Z}\right]\,{\rm{sin}}\omega (t - {t_6})\nonumber\\ &&+\, \left[\frac{{{i_l}_m({t_6})}}{{n + 1}} + {i_{lk}}({t_6})\right]{\rm{cos}}\omega (t - {t_6}) \end{eqnarray}19Vc1(t)=Vo−2Vc1+nV1n+1+2Vc1−Vo+nV1n+1+Cs3Cs1+Cs3Vcc+Cs1Cs1+Cs3Vc3(t6)cosω(t−t6)+ilm(t6)n+1Z+ilk(t6)(n+1)ilm(t6)n+1sinω(t−t6)\begin{eqnarray} {V_{c1}}(t) &=& \dfrac{{{V_o} - 2\,{V_{c1}} + n\,{V_1}}}{{n + 1}} + \left[\dfrac{{2\,{V_{c1}} - {V_o} + n\,{V_1}}}{{n + 1}}+ \dfrac{{{C_{s3}}}}{{{C_{s1}} + {C_{s3}}}}{V_{cc}}\right.\nonumber\\ &&+\, \left.\dfrac{{{C_{s1}}}}{{{C_{s1}} + {C_{s3}}}}{V_{c3}}({t_6})\right]\,{\rm{cos}}\omega (t - {t_6}) + \left[\dfrac{{{i_{lm}}({t_6})}}{{n + 1}}Z\right.\nonumber\\ &&+\,\left. {i_{lk}}({t_6})\,(n + 1)\vphantom{\dfrac{{{i_{lm}}({t_6})}}{{n + 1}}}\right]\,{\rm{sin}}\omega (t - {t_6}) \end{eqnarray}In this mode, resonance impedance (Zo) and angular resonance frequency (ωo) are equal to:20Z0=LLK(Cs1+Cs3)\begin{equation}{Z_0} = \sqrt {\frac{{{L_{LK}}}}{{({C_{s1}} + {C_{s3}})}}} \end{equation}21ω0=n+1n(Cs1+Cs3)LLK\begin{equation}{\omega _0} = \frac{{n + 1}}{{n\,\sqrt {({C_{s1}} + {C_{s3}})\,{L_{LK}}} }}\end{equation}Mode 8—Time range (t7 – t8)The capacitor voltage of the main switch VCs1 reaches zero at the end of this mode and the body diode begins to conduct. The equations for this mode of operation are calculated as below:22ilm(t)=ilm(t7)+2Vc1−Vonlm(t−t7)\begin{equation}{i_{lm}}(t) = {i_{lm}}({t_7}) + \frac{{2\,{V_{c1}} - {V_o}}}{{n\,{l_m}}}(t - {t_7})\end{equation}23ilk(t)+Vin−Vlmllk(t−t0)=ilk(t7)+Vin−(2Vc1−Von)llk(t−t7)\begin{equation}{i_{lk}}(t) + \frac{{{V_{in}} - {V_{lm}}}}{{{l_{lk}}}}(t - {t_0}) = {i_{lk}}({t_7})+ \frac{{{V_{in}} - (\frac{{2\,{V_{c1}} - {V_o}}}{n})}}{{{l_{lk}}}}(t - {t_7})\end{equation}ESS charging modeIn ESS charging mode, the power generated by the upper source is much more than the required power of the load, and if ESS charging is required, additional energy is used to charge the ESS. In this mode, the proposed converter acts similarly to a buck converter and charging the ESS through the lower phase, and so the upper phase acting as the output load supplier. switching is done in the form of hard switching in this mode, because the converter performs as a buck converter and the buck converter by itself has an efficiency of over 90%. So hard switching is used, which does not complicate the circuit and has a proper efficiency in this mode. In this mode, S2 switch is always off and S4 acts as the main converter switch. Figure 6 shows the main waveform diagrams of the proposed converter in ESS charging mode. According to Figure 7, the proposed converter in ESS charging mode has 2 operating modes in one switching period.6FIGUREMain key waveform diagrams in the proposed converter in ESS charging mode7FIGUREOperating modes in ESS charging mode of proposed converterMode 1—Time range (t0 – t1)The time range in this mode is between t0 and t1. This mode starts where the S4 is still on. The current of iLm2 becomes positive and its value begins to increase. The equivalent circuit of this mode is shown in Figure 7a. In this case the following equations are established:24Vcc2=V1\begin{equation}{V_{cc2}} = {V_1}\end{equation}25VLm2=V1−V2\begin{equation}{V_{Lm2}} = {V_1}\; - {V_2}\end{equation}26iLm2t=iLm2t0+VLm2Lm2t−t0\begin{equation}{i_{Lm2}}\left( t \right)\; = \;{i_{Lm2}}\left( {{t_0}} \right) + \frac{{{V_{Lm2}}}}{{{L_{m2}}}}\left( {t - {t_0}} \right)\end{equation}27iLm2t=iLm2t0+V2−V1lm2t−t0\begin{equation}{i_{Lm2}}\left( t \right) = {i_{Lm2}}\left( {{t_0}} \right) + \frac{{{V_2} - {V_1}}}{{{l_{m2}}}}\left( {t - {t_0}} \right)\end{equation}Mode 2—Time range (t1 – t2)Time range in this mode is between t1 and t2. This mode starts where the S4 is on. Similar to the previous mode the current of iLm2 becomes positive and increase. The equivalent circuit of this mode is shown in Figure 7b. The equations for this operation mode are calculated as bellow:28VLm2=V2\begin{equation}{V_{Lm2}} = {V_2}\end{equation}29iLm2t=iLm2t1+VLm2Lm2t−t1\begin{equation}{i_{Lm2}}\left( t \right) = {i_{Lm2}}\left( {{t_1}} \right) + \frac{{{V_{Lm2}}}}{{{L_{m2}}}}\left( {t - {t_1}} \right)\end{equation}30iLm2t=iLm2t1+V2lm2t−t1\begin{equation}{i_{Lm2}}\left( t \right) = {i_{Lm2}}\left( {{t_1}} \right) + \frac{{{V_2}}}{{{l_{m2}}}}\left( {t - {t_1}} \right)\end{equation}HIGH STEP‐UP MULTI‐INPUT DC‐DC PROPOSED CONVERTER DESIGNIn ESS discharging modeThe design method along with the analysis of the proposed converter are presented in the following four sections. For the simplicity of the equations, all the elements of the circuit are ideally considered.Voltage gainWhen the main switch of each phase is on, the inductor Lm is charged by the input voltage Vin and the inductor voltage in the time interval from zero to DT is equal to:31Vlm=Vin\begin{equation}{V_{lm}} = {V_{in}}\end{equation}When the main switch is turned off, the inductor Lm starts to discharge, the voltage value of which is calculated in the time interval DT to 1‐DT as follows:32Vlm=(2n+1)Vin−Von+1\begin{equation}{V_{lm}} = \frac{{{\rm{(2}}n + {\rm{1)}}\,{V_{in}} - {V_o}}}{{n + {\rm{1}}}}\end{equation}Using the volt‐second balance, the voltage gain is given by:33VoVin=(2n+1)−nD1−D\begin{equation}\frac{{{V_o}}}{{{V_{in}}}} = \frac{{(2n + 1)\, - nD}}{{1 - D}}\end{equation}Voltage stressVoltage stress is calculated when the switch is off. Voltage stress of the main and auxiliary switches in the off state is equal to the voltage of the clamp capacitor.34Vsw3=Vsw1=Vcc=−nVin+Von+1\begin{equation}{V_{sw3}} = {V_{sw1}} = {V_{cc}} = - \frac{{n{V_{in}} + \,{V_o}}}{{n + 1}}\end{equation}The stress voltage of the output diode when the diode is off being calculated as bellow:35Vlm=nVin\begin{equation}{V_{lm}} = n{V_{in}}\end{equation}36VDo1=Vo−nVin\begin{equation}{V_{Do1}} = {V_o} - n{V_{in}}\end{equation}The voltage stress of diodes Dd3 and Dd1 is given by:37Vc1=Vc2=nVin\begin{equation}{V_{c1}} = {V_{c2}} = n{V_{in}}\end{equation}38Vlm=VL1\begin{equation}{V_{lm}} = V{L_1}\end{equation}39Vl2=nVlm\begin{equation}{V_{l2}} = n{V_{lm}}\end{equation}40VDd3=VDd1=−Vc1+V12=nVin−Vo\begin{equation}{V_{Dd3}} = {V_{Dd1}} = - \,{V_{c1}} + {V_{12}} = n\,{V_{in}} - {V_o}\end{equation}Current stressThe switch current stress is in fact the maximum current that passes through when the switch is off. It is calculated as follows:41Ismax=2(1+2n)−nDD(1−D)Io\begin{equation}{I_{s\max }} = 2\left[\frac{{(1 + 2\,n) - n\,D}}{{D\,(1 - D)}}\right]\,{I_o}\end{equation}The current stress of the output diode and diodes Dd3 and Dd1 are equal to:42IDomax=2Io1−D\begin{equation}{I_{Do\,max }} = \frac{{2\,{I_o}}}{{1 - D}}\end{equation}43IDd1max=IDd3max=2IoD\begin{equation}{I_{Dd1max}} = {I_{Dd3\,max }} = \frac{{2\,{I_o}}}{D}\end{equation}Design of the proposed converter elementsIn this section, the equations of the proposed converter elements, including output capacitors and coupled inductor, are described.Coupled inductor designIn most of the time, it is better to select the value of the inductor so that the converter has 20% of the current ripple on the inductor at the maximum amount of output power. Figure 8a shows the boundary conduction mode (BCM) and Figure 8b shows the inductor current waveform in continuous conduction mode (CCM). The minimum amount of magnetic inductance required to hold the converter in CCM mode can be expressed as bellow.44ΔIlm=20%Ilm\begin{equation}\Delta {I_{lm}} = 20\% \,{I_{lm}}\end{equation}45ΔIlm=2Ilm\begin{equation}\Delta {I_{lm}} = 2\,{I_{lm}}\end{equation}8FIGUREInductor current waveform. (a) In CCM mode and (b) in BCM modeMaximum load resistanceThe maximum load resistance (RlBCM) can be selected at about 20% of the nominal output power, in other words, it can be said to be five times the amount of output load. If p1 and p2 are upper phase power and lower phase power, respectively, the total power value (p) is equal to:46p=p1+p2\begin{equation}p = {p_1} + {p_2}\end{equation}47p=vo2RL\begin{equation}p = \frac{{v_o^2}}{{{R_L}}}\end{equation}48RlBCM=5RL\begin{equation}{R_{lBCM}} = 5\,{R_L}\end{equation}According to the nominal output power, the converter equivalent circuit and the average inductor current waveform in BCM mode are shown in Figures 9 and 10, respectively. The average inductor current is calculated as follows:49Ilm=Ilm(1−D)→Ilm=(n+1)Io(1−D)\begin{equation}{I_{lm}} = {I_{lm}}(1 - D) \to {I_{lm}} = \frac{{(n + 1)\,{I_o}}}{{(1 - D)}}\end{equation}50ΔIlm=vlmDTLm\begin{equation}\Delta {I_{lm}} = \frac{{{v_{lm}}D\,T}}{{{L_m}}}\end{equation}9FIGUREConverter equivalent circuit in BCM mode10FIGUREAverage inductor current waveform in BCM modeBy placing the above equations, the value of the inductor Lm is equal to:51Lm=D(1−D)2Rlbcm2(n+1)[(1+2n)−nD]fs\begin{equation}{L_m} = \frac{{D\,{{(1 - D)}^2}\,{R_{lbcm}}}}{{2\,(n + 1)\,[(1 + 2n) - nD]\,{f_s}}}\end{equation}Output capacitor designThe value of the output capacitor is basically selected according to the output voltage ripple. The legal output voltage ripple is considered to be 0.1% of the output voltage. due to the average current passing through the capacitor is zero, the output capacitor value is calculated according to the following equation:52Co≥Io·DΔVCo·fs\begin{equation}{C_o} \ge \frac{{{I_o}\cdot D}}{{\Delta {V_{Co}}\cdot {f_s}}}\end{equation}The value of the switch capacitor fundamentally depends on the voltage ripple and output power level. The values of capacitors C1 and C2 are determined from the following equation:53C1≥Io1−DΔVC1·fs\begin{equation}\;{C_1} \ge \frac{{{I_o}\left( {1 - D} \right)}}{{\Delta {V_{C1}}\cdot {f_s}}}\end{equation}The values of capacitors C1 and C2 are determined from the following equation:54Cc1≥VoΔVCc1·fs·RL\begin{equation}{C_{c1}} \ge \frac{{{V_o}}}{{\Delta {V_{{C_{c1}}}}\cdot {f_s}\cdot {R_L}}}\end{equation}Soft switchingIn power electronic converters, soft switching techniques are used to reduce losses by minimizing the overlap of the current and voltage of the switch during its on and off times. This paper uses the ZVS technique. Also the resonance phenomenon is used to zero the voltage across the two switches. It is clear that the turn‐off the switches conditions under ZVS are provided by snubber capacitors parallel to the switches. before switching, reverse current enters the body diode of the transistor and discharges the snubber capacitor as a result, the voltage across the two heads approaches zero, which provides the turn‐on conditions under ZVS. In this situation, the amount of energy stored in the leakage inductance must be more than the energy stored in the capacitor parallel to the switches:5512LLKiLK2(tx)≥12CsVds2(tx)\begin{equation}\frac{1}{2}{L_{LK}}i_{LK}^2({t_x}) \ge \,\frac{1}{2}{C_s}V_{ds}^2({t_x})\end{equation}5612LLK(n+1)Io1−D≥Cs1||Cs3vin1−D2\begin{equation}\frac{1}{2}{L_{LK}}\left[\frac{{(n + 1)\,{I_o}}}{{1 - D}}\right] \ge {C_{s1}}||\,{C_{s3}}\,{\left(\frac{{{v_{in}}}}{{1 - D}}\right)^2}\end{equation}According to the amount of load stored in the capacitors and their discharge current, the amount of dead time is calculated from the following equation.57Td>(c2+c4)V1ILmin\begin{equation}{T_d} > \,\frac{{({c_2} + {c_4})\,{V_1}}}{{{I_{L\min }}}}\end{equation}In ESS charging modeAs seen in the operating modes of the proposed converter in ESS charging mode, in this mode the converter works exactly the same as a buck converter, which includes elements of switches S2 and S4 and source V1 and diode D as well as inductor Lm2. The V2 storage source is charged by a buck converter. Given that in this operating mode the proposed converter performs similarly to a buck converter; the voltage gain as well as all the equations are exactly the same as the buck converter equations.SIMULATION RESULTSIn this section, the simulation results of the proposed converter using ORCOD software in different performance modes to confirm the theoretical analysis of the proposed converter are shown. The parameters of the proposed converter are given in Table 1. The simulation results are a function of the time shown and the time axis is in milliseconds.1TABLEParameter values in the proposed converterComponentsValuePower400 WFirst input voltage40 VSecond input voltage24 VOutput voltage400 VSwitching frequency100 kHzOutput capacitors38 μFSwitched capacitor4.7 μFClamp capacitor4.7 μFSwitchesIRFP260DiodesMUR840 and MUR460Simulation results in ESS discharge modeInput voltages are the main parameters for the design of the buck converter inductor to operate in BCM mode and provide soft switching conditions. Assuming the first input voltages (V1) and the second (V2) are 40 and 24 V, respectively, and the duty cycle (D) is greater than or equal to 0.5, and the value of the turns ratio for upper‐phase coupled inductors (n1) is 2.14. And for lower coupled inductor, (n2) is equal to 4.04. According to the output power and accurate calculation of RlBCM, the values of magnetic inductors in the upper (Lm1) and lower (Lm2) phases are equal to 150 and 57 μH, respectively. Leakage inductor and magnetic inductor current in the upper and lower phases are shown in Figures 11 and 12. As can be seen, the ripple magnetic inductor current is low, therefore the ohmic losses of the inductor and the switches are reduced, and as shown in the simulation results, the leakage inductance current at certain time intervals in the second section shows the negative values and when occurs the amount of iLK current becomes less than iLm and the direction of the current changes (the polarity of the mating inductor changes). Figure 13 and 14, show the current waveforms of the voltage multiplier capacitor and the clamp capacitor in the upper and lower phases, respectively. Figure 15, also shows the waveform of the output diode current in the upper and lower phases and shows the operating modes of the output diode correctly. As can be seen, at intervals when the auxiliary switch S3 is turned on under ZVS conditions, the current passes through the diode as long as the current of the switch is positive, and as soon as it is negative, the current will pass through the switch.11FIGURELeakage inductor current and magnetic inductor in upper phase12FIGURELeakage inductor current and magnetic inductor in lower phase13FIGUREVoltage switch‐capacitor current waveform in upper and lower phase14FIGUREClamp capacitor current waveform in upper and lower phase15FIGUREOutput diode current waveform in upper and lower phaseFigures 16 and 17 show the soft switching condition of the main and auxiliary switches of the upper and lower phases. As can be seen, the overlap of the current and voltage of the switch has reached its minimum value when it is turned on and off. Figure 18 shows the input current waveform from the 40 and 24 V source, respectively.16FIGURESoft switching condition of main keys (S1 and S2)17FIGURESoft switching condition of auxiliary keys (S3 and S4)18FIGUREInput current waveform 40 and 24 V sourcesSimulation results in ESS charging modeIn ESS charging mode, the input voltages are the main parameters for the design of the buck converter inductor to operate in BCM mode. Assuming the first input voltages (V1) and the second (V2) are 40 and 24 V, respectively, and the duty cycle (D) is greater than or equal to 0.5, and the value of the turns ratio for upper‐phase coupled inductors (n1) is 2.14. And for the lower phase coupled inductor (n2) is equal to 4.04. All values related to the design of the inductor and etc are similar to the discharge mode. The results of the simulation at full load are shown in Figures 19–27. Leakage inductance current in ESS charge mode is shown in Figure 19. gate‐source and drain‐source voltages on the switch S4 are given in Figures 20 and 21, respectively. The switch current waveform is shown in Figure 22. Gate‐source and drain‐source voltages on the switch S2 are shown in Figures 23 and 24, respectively. As can be seen, gate‐source voltage at switch S2 is zero, and in fact this switch is off. The switch S2 current waveform of can be seen in Figure 25. The low phase output diode waveform shown in Figure 26 display that no current is transmitted from the S2 switch to the output. Figure 27 shows the input current waveform of the source 40 and 24 V, respectively.19FIGUREInductor current waveform in ESS charging mode20FIGUREGate‐source voltage waveform of switch S4 in ESS charging mode21FIGUREDrain‐source voltage waveform of switch S4 in ESS charging mode22FIGURESwitch S4 current waveform in ESS charging mode23FIGUREGate‐source voltage waveform of switch S2 in ESS charging mode24FIGUREDrain‐source voltage waveform of switch S2 in ESS charging mode25FIGURESwitch S2 current waveform in ESS charging mode26FIGUREDiode output current waveform in lower phase in ESS charging mode27FIGUREInput current waveform of source 40 and 24 V in ESS charging modeA comparison between several similar converters with the proposed converter is given in Table 2. As can be seen, the proposed converter has a higher efficiency as well as higher output power compared to similar converters. As a result, in high step‐up applications, it can be used more than similar converters. In three converters, soft switching under zero voltage switching (ZVS) and in one converter, soft switching under zero current switching (ZCS) is used.2TABLEComparison of the proposed converter with similar convertersConverter topology[47][48][49][50][51]Proposed converterNumber of switch224224Number of Diode335627Voltage gainDNt+Ns+21−D$\frac{{D{N_t} + {N_s} + 2}}{{1 - D}}$1+2N−ND1−D$\frac{{1 + 2N - ND}}{{1 - D}}$1+N1−D$\frac{{1 + N}}{{1 - D}}$2+2N1−D$\frac{{2 + 2N}}{{1 - D}}$2+N1−D$\frac{{2 + N}}{{1 - D}}$(2N+1)−ND1−D$\frac{{( {2N + 1} ) - ND}}{{1 - D}}$Voltage stress on switchVs1−D$\frac{{{V_s}}}{{1 - D}}$Vo−NVin1+N$\frac{{{V_o} - N{V_{in}}}}{{1 + N}}$VoN+1$\frac{{{V_o}}}{{N + 1}}$Vo2(N+1)$\frac{{{V_o}}}{{2( {N + 1} )}}$Vo2+N$\frac{{{V_o}}}{{2 + N}}$−NVin+VoN+1$ - \frac{{N{V_{in}} + {V_o}}}{{N + 1}}$Voltage stress on output diodeDNt1−DVg$\frac{{D{N_t}}}{{1 - D}}{V_g}$(1+N)Vo1+2N−ND$\frac{{( {1 + N} ){V_o}}}{{1 + 2N - ND}}$Vo${V_o}$NVo(N+1)$\frac{{N{V_o}}}{{( {N + 1} )}}$(1+N)Vo2+N$\frac{{( {1 + N} ){V_o}}}{{2 + N}}$Vo−nVin${V_o} - n{V_{in}}$Multi‐input/port××√××√Soft switchingZVSZVSZVSZCSZVSZVSEnergy storage capability××√××√Method to reduce no. of componentsSoft switching techniquesActive clamp and combination of coupled inductor and switched capacitorIntegrating ESS charging and discharging paths with each otherCombines the advantages of switched capacitors, coupling inductors, and voltage multiplier techniquesZVS soft switching techniquesIntegrating ESS charging and discharging paths with each other and combination of the inductance coupling method with the voltage multiplierEfficiency96.29595.3394.794.897.63EXPERIMENTAL RESULTIn this section, the practical results obtained from the proposed converter in different operating modes are shown to confirm the theoretical analysis of the proposed converter. The laboratory sample structure of the proposed converter and how it is connected to the scope are shown in Figures 28 and 29.28FIGURELaboratory sample of the proposed converter29FIGUREConnecting the proposed converter to the load and scopeExperimental results in ESS discharging modeFigure 30 shows the soft switching condition in the main keys and Figure 31 shows the soft switching condition in the S3 and S4 keys in the ESS discharge mode. Figures 32 and 33 show the input current waveform of source 40 and 24 V, respectively.30FIGURESoft switching condition on S1 and S2 keys in ESS discharge mode31FIGURESoft switching condition on S1 and S2 keys in ESS discharge mode32FIGUREInput current waveform of source 4033FIGUREInput current waveform of source 24Experimental results in ESS charging modeFigure 34 shows the condition of the S2 switch in ESS charge mode. As can be seen, the S2 switch is off in this mode and only its diode conducts. Figure 35 shows the condition of the S4 key in ESS charge mode. Figure 36 and 37 show the input current waveform of source 40 and 24 V, respectively.34FIGURES2 key condition in ESS charging mode35FIGURES4 key condition in ESS charging mode36FIGUREInput current waveform of source 40 in ESS charging mode37FIGUREInput current waveform of source 40 in ESS charging modeAs the results obtained from the construction of the proposed converter show, the practical results completely confirm the simulation results. Figure 38 and 39 show the proposed converter efficiency under different output load conditions. As can be seen, in most load conditions, the proposed converter efficiency in the ESS discharge mode remains in the range of 97%. The maximum measured efficiency of this converter is 97.63%. In ESS charging mode, the proposed converter efficiency remains in the range of 95% and its maximum value is 95.37%. According to the results obtained from the simulation of the converter as well as the practical results obtained from the construction of the proposed converter, this converter has a suitable increase voltage gain and it has the highest possible efficiency in a range of loads and also all switches operate in all operating modes under soft switching conditions.38FIGUREThe efficiency of the proposed converter in different load conditions in ESS discharging mode39FIGUREThe efficiency of the proposed converter in different load conditions in ESS charging modeThe loss analysis of the proposed converter at full load is shown in Table 3 is given, which confirms the correctness of the proposed converter efficiency.3TABLEProposed converter analysisParametersLosses (MW)Do1543.37Do2818.24Dd1620.83Dd2645.69Dd3620.83Dd4645.69S12.92 × 103S25.8 × 103S3807.5S40.998 × 103According to the results obtained from the simulation of the proposed converter, some of the features and advantages of this converter can be described as follows:(a)Reduce the volume of the circuit.(b)Creating a soft switching condition for the active circuit elements, which has the ability to increase the switching frequency and reduce the volume of the converter.(c)Non‐isolated (same input and output ground, simple structure, no need for isolated transformer and low construction cost).(d)Efficiency increase of about 97.63%.CONCLUSIONIn this paper a high step‐up non‐isolated multi‐input converter is proposed, as we know, renewable energy sources are inherently low voltage, so high step‐up techniques are used to increase their voltage gain. Due to the non‐isolation of the proposed converter (the same input and output ground, no need for isolated transformers and also simple structure), the construction cost is lower compared to similar converters. Also, creating soft switching conditions for active circuit elements has increased the switching frequency and reduced the converter volume. This converter has the ability to connect to other high‐step‐up converters, which include the coupled inductor and the active clamp structure, and converts them into multi‐input converters. According to the simulation results, the proposed converter has an increase in the appropriate voltage gain and has the highest possible efficiency in a load range. Precisely the maximum measured efficiency of this converter is 97.63% and in most load conditions, the efficiency remains in the range of 97%, and also, all switches work in all operating modes under soft switching conditions. In order to confirm the results of the simulation of the proposed converter, a laboratory sample of the proposed converter was made. The practical results obtained from the proposed converter in different operating modes were presented to confirm the theoretical analysis of the proposed converter and the accuracy of construction was shown.FUNDING INFORMATIONThere are no funders to report for this submission.CONFLICT OF INTERESTThe authors declare that they have no known competing financial interests or personal relationships that could have appeared to influence the work reported in this paper.DATA AVAILABILITY STATEMENTData openly available in a public repository that issues datasets with DOIsREFERENCESFaiz, J., Shahgholian, G., Ehsan, M.: Stability analysis and simulation of a single‐phase voltage source UPS inverter with two‐stage cascade output filter. Eur. Trans. Electr. Power 18(1), 29–49 (2008). https://doi.org/10.1002/etep.160Borhani, M., Yaghoubi, S.: Improvement of energy dissipative particle dynamics method to increase accuracy. J. Therm. Anal. Calorim. 144, 2543–2555 (2021). https://doi.org/10.1007/s10973‐020‐10362‐1Shahgholian, G.: A brief review on microgrids: Operation, applications, modeling, and control. Int. Trans. Electr. Energy Syst. 31, e12885 (2021). https://doi.org/10.1002/2050‐7038.12885Niazvand, F., Kharrati, S., Khosravi, F., Rastgou, A.: Dual‐objectives energy and load management for an energy hub by considering diverse plannings and in the presence of CCUS technology and the TOU program. J. Intell. Proced. Electr. Technol. 14(54), 31–58 (2023). https://dorl.net/dor/20.1001.1.23223871.1402.14.54.3.9Shahgholian, G., Shafaghi, P.: State space modeling and eigenvalue analysis of the permanent magnet DC motor drive system. In: Proceeding of the IEEE/ICECT. Kuala Lumpur, Malaysia, pp. 63–67 (2010). https://doi.org/10.1109/ICECTECH.2010.5479987Kharrazi, A., Sreeram, V., Mishra, Y.: Assessment techniques of the impact of grid‐tied rooftop photovoltaic generation on the power quality of low voltage distribution network‐ A review. Renewable Sustainable Energy Rev. 120, 109643 (2020). https://doi.org/10.1016/j.rser.2019.109643Alam, M., Kumar, K., Dutta, V.: Implementation of multiple outputs fly back converter for dc microgrid application. In: Proceeding of the IEEE/ICESIP. Chennai, India, pp. 1–4 (2019). https://doi.org/10.1109/ICESIP46348.2019.8938288Jahanshahi Bavandpour, R., Ghadiri, H., Khodadadi, H.: Optimal design of a hybrid solar–wind–battery system using the grasshopper optimization algorithm for minimization of the loss of power supply probability. J. Intell. Proced. Electr. Technol. 13(51), 139–156 (2022). https://dorl.net/dor/20.1001.1.23223871.1401.13.51.9.2Yi, W., Ma, H., Peng, S., Liu, D., Ali, Z.M., Dampage, U., Hajjiah, A.: Analysis and implementation of multi‐port bidirectional converter for hybrid energy systems. Energy Rep. 8, 1538–1549 (2022). https://doi.org/10.1016/j.egyr.2021.12.068Xu, J., Liu, L., Wang, F.: Equilibrium strategy‐based economic‐reliable approach for day‐ahead scheduling towards solar‐wind‐gas hybrid power generation system: A case study from China. Energy. 240, 122728 (2022). https://doi.org/10.1016/j.energy.2021.122728Pal, S., Sinha, D.: Design and simulation of modified multi‐input interleaved boost converter for renewable resources. In: Proceeding of the IEEE/IEMRE. Kolkata, India, pp.1–6 (2021). https://doi.org/10.1109/IEMRE52042.2021.9386869Asy'ari, M.K., Musyafa, A.’, Indriawati, K.: Design of wind turbine output voltage control systems in multi‐input buck converter using fuzzy logic control for battery charging. In: Proceeding of the IEEE/ICAMIMIA). Batu, Malang, Indonesia, pp. 249–252 (2019). https://doi.org/10.1109/ICAMIMIA47173.2019.9223417Haghshenas, G., Mirtalaei, S.M.M., Mordmand, H., Shahgholian, G.: High step‐up boost‐flyback converter with soft switching for photovoltaic applications. J. Circuits, Syst. Comput. 28(1), 1–16 (2019). https://doi.org/10.1142/S0218126619500142Fernandez, A.M., Kandidayeni, M., Boulon, L., Chaoui, H.: An adaptive state machine based energy management strategy for a multi‐stack fuel cell hybrid electric vehicle. IEEE Trans. Veh. Technol. 69(1), 220–234 (2020). https://doi.org/10.1109/TVT.2019.2950558Güler, N., Irmak, E.: Design, implementation and model predictive based control of a mode‐changeable DC/DC converter for hybrid renewable energy systems. ISA Trans. 114, 485–498 (2021). https://doi.org/10.1016/j.isatra.2020.12.023Vidhya, S.D., Balaji, M.: Hybrid fuzzy PI controlled multi‐input DC/DC converter for electric vehicle application. Automatika 61(1), 79–91 (2020). https://doi.org/10.1080/00051144.2019.1684038Affam, A., Buswig, Y.M., Othman, A.K.B.H., Julai, N.B., Qays, O.: A review of multiple input DC‐DC converter topologies linked with hybrid electric vehicles and renewable energy systems. Renewable Sustainable Energy Rev. 135, 110186 (2021). https://doi.org/10.1016/j.rser.2020.110186Sharifiyana, O., Dehghani, M., Shahgholian, G., Mirtalaei, S.M.M., Jabbari, M.: An overview of the structure and improvement of the main parameters of non‐isolated dc/dc boost converters. J. Intell. Proced. Electr. Technol. 12(48), 1–29 (2022). https://dorl.net/dor/20.1001.1.23223871.1400.12.48.6.6Lee, S., Kang, B.: Quasi‐resonant passive snubber for improving power convers ion efficiency of a dc–dc step‐down converter. IEEE Trans. Power Electron. 33, 2026–2034 (2018). https://doi.org/10.1109/TPEL.2017.2698211Rajan, P., Jeevananthan, S.: A new partially isolated hybrid output of multiport multilevel converter for photovoltaic based power supplies. J. Energy Storage 45, 103436 (2022). https://doi.org/10.1016/j.est.2021.103436Revathi, B.S., Mahalingam, P., Gonzalez‐Longatt, F.: Interleaved high gain DC‐DC converter for integrating solar PV source to DC bus. Sol. Energy. 188, 924–934 (2019). https://doi.org/10.1016/j.solener.2019.06.072Reddi, N.K., Ramteke, M.R., Suryawanshi, H.M., Kothapalli, K., Gawande, S.P.: An isolated multi‐input ZCS DC–DC front‐end‐converter based multilevel inverter for the integration of renewable energy sources. IEEE Trans. Ind. Appl. 54(1), 494–504 (2018). https://doi.org/10.1109/TIA.2017.2753160Nawaz, A., Wu, J., Long, C.: Mitigation of circulating currents for proportional current sharing and voltage stability of isolated DC microgrid. Electr. Power Syst. Res. 180, 106123 (2020). https://doi.org/10.1016/j.epsr.2019.106123Rani, P.H., Navasree, S., George, S., Ashok, S.: Fuzzy logic supervisory controller for multi‐input non‐isolated DC to DC converter connected to DC grid. Int. J. Electr. Power Energy Syst. 112, 49–60 (2019). https://doi.org/10.1016/j.ijepes.2019.04.018Amaleswari, R., Prabhakar, M.: Non‐isolated multi‐input dc‐dc converter with current sharing mechanism. Int. J. Electron. 108(2), 237–263 (2021). https://doi.org/10.1080/00207217.2020.1789760Reddy, K.J., Natarajan, S.: Energy sources and multi‐input dc‐dc converters used in hybrid electric vehicle applications– A review. Int. J. Hydrogen Energy. 43(36), 17387–17408 (2018). https://doi.org/10.1016/j.ijhydene.2018.07.076Zhang, N., Sutanto, D., Muttaqi, K.M.: A review of topologies of three‐port dc–dc converters for the integration of renewable energy and energy storage system. Renewable Sustainable Energy Rev. 56), 388–401 (2016). https://doi.org/10.1016/j.rser.2015.11.079Vidhya, S.D., Balaji, M.: Hybrid fuzzy PI controlled multi‐input DC/DC converter for electric vehicle application. Automatika. 61(1), 79–91 (2020). https://doi.org/10.1080/00051144.2019.1684038Affam, A., Buswig, Y.M., Othman, A.B.H., Julai, N.B., Qays, O.: A review of multiple input DC‐DC converter topologies linked with hybrid electric vehicles and renewable energy systems. Renewable Sustainable Energy Rev. 135, 110186 (2021). https://doi.org/10.1016/j.rser.2020.110186Liu, Y., Chen, Y.: A systematic approach to synthesizing multi‐input DC–DC converters. IEEE Trans. Power Electron. 24(1), 116–127 (2009). https://doi.org/10.1109/TPEL.2008.2009170Bairabathina, S., Balamurugan, S.: Review on non‐isolated multi‐input step‐up converters for grid‐independent hybrid electric vehicles. Int. J. Hydrogen Energy. 45(41), 21687–21713 (2020). https://doi.org/10.1016/j.ijhydene.2020.05.277Chen, G., Liu, Y., Qing, X., Wang, F.: Synthesis of integrated multiport dc–dc converters with reduced switches. IEEE Trans. Ind. Electron. 67(6), 4536–4546 (2020). https://doi.org/10.1109/TIE.2019.2931214Shan, Z., Ding, X., Jatskevich, J., Tse, C.K.: Synthesis of multi‐input multi‐output dc/dc converters without energy buffer stages. IEEE Trans. Circuits Syst. II: Express Briefs. 68(2), 712–716 (2021). https://doi.org/10.1109/TIE.2019.2931214Mohammadi, M.R., Farzanehfard, H.: Family of soft‐switching bidirectional converters with extended ZVS range. IEEE Trans. Ind. Electron. 64(9), 7000–7008 (2017). https://doi.org/10.1109/TIE.2017.2686308Elserougi, A., Abdelsalam, I., Massoud, A., Ahmed, S.: A bidirectional non‐isolated hybrid modular DC–DC converter with zero‐voltage switching. Electr. Power Syst. Res. 167, 277–289 (2019). https://doi.org/10.1016/J.EPSR.2018.11.009Kanathipan, K., Moury, S., Lam, J.: A fast and accurate maximum power point tracker for a multi‐input converter with wide range of soft‐switching operation for solar energy systems. In: Proceeding of the IEEE/APEC. Tampa, FL, pp. 2076–2083 (2017). https://doi.org/10.1109/APEC.2017.7930985Kwon, J.M., Kwon, B.H.: High step‐up active‐clamp converter with input‐current doubler and output‐voltage doubler for fuel cell power systems. IEEE Trans. Power Electron. 24(1), 108–115 (2009). https://doi.org/10.1109/TPEL.2008.2006268Faraji, R., Ding, L., Rahimi, T., Kheshti, M., Islam, M.R.: Soft‐switched three‐port dc‐dc converter with simple auxiliary circuit. IEEE Access. 9, 66738–66750 (2021). https://doi.org/10.1109/ACCESS.2021.3076183Salehi, N., Mirtalaei, S.M.M., Mirenayat, S.H.: A high step‐up dc–dc soft‐switched converter using coupled inductor and switched capacitor. Int. J. Electron. Lett. 6(3), 260–271 (2018). https://doi.org/10.1080/21681724.2017.1357195Reddi, N.K., Ramteke, M.R., Suryawanshi, H.M., Kothapalli, K., Gawande, S.P.: An isolated multi‐input ZCS DC–DC front‐end‐converter based multilevel inverter for the integration of renewable energy sources. IEEE Trans. Ind. Appl. 54(1), 494–504 (2018). https://doi.org/10.1109/TIA.2017.2753160Li, W., He, X.: Review of nonisolated high‐step‐Up DC/DC converters in photovoltaic grid‐connected applications. IEEE Trans. Ind. Electron. 58(4), 1239–1250 (2011). https://doi.org/10.1109/TIE.2010.2049715Andrade, J.M., Coelho, R.F., Lazzarin, T.B.: High step‐up dc–dc converter based on modified active switched‐inductor and switched‐capacitor cells. IET Power Electron. 13(14), 3127–3137 (2020). https://doi.org/10.1049/iet‐pel.2020.0064Waseem, M., Saeed, L., Khan, M.Y.A., Saleem, J., Majid, A.: A multi input multi output bidirectional DC‐DC boost converter with backup battery port. In: Proceeding of the IEEE/ICPESG. Mirpur Azad Kashmir, Pakistan, pp. 1–6 (2018). https://doi.org/10.1109/ICPESG.2018.8384526Revathi, B.S., Mahalingam, P., Gonzalez‐Longatt, F.: Interleaved high gain dc‐dc converter for integrating solar PV source to dc bus. Sol. Energy 188, 924–934 (2019). https://doi.org/10.1016/j.solener.2019.06.072Schmitz, L., Martins, D.C., Coelho, R.F.: Comprehensive conception of high step‐up dc–dc converters with coupled inductor and voltage multipliers techniques. IEEE Trans. Circuits Syst. I: Regul. Pap. 67(6), 2140–2151 (2020). https://doi.org/10.1109/TCSI.2020.2973154Ye, Y., Chen, S., Yi, Y.: Switched‐capacitor and coupled‐inductor‐based high step‐up converter with improved voltage gain. IEEE J. Emerging Sel. Top. Power Electron. 9(1), 754–764 (2021). https://doi.org/10.1109/JESTPE.2020.2971525Zhang, X., et al.: Novel high step‐up soft‐switching dc–dc converter based on switched capacitor and coupled inductor. IEEE Trans. Power Electron. 35(9), 9471–9481 (2020). https://doi.org/10.1109/TPEL.2020.2972583Sathyan, S., Suryawanshi, H.M., Ballal, M.S., Shitole, A.B.: Soft‐switching dc–dc converter for distributed energy sources with high step‐up voltage capability. IEEE Trans. Ind. Electron. 62(11), 7039–7050 (2015). https://doi.org/10.1109/TIE.2015.2448515Faraji, R., Farzanehfard, H.: Soft‐switched nonisolated high step‐up three‐port dc–dc converter for hybrid energy systems. IEEE Trans. Power Electron. 33(12), 10101–10111 (2018). https://doi.org/10.1109/TPEL.2018.2791840Seo, S.W., Ryu, J.H., Kim, Y., Choi, H.H.: Non‐isolated high step‐up dc/dc converter with coupled inductor and switched capacitor. IEEE Access. 8, 217108–217122 (2020). https://doi.org/10.1109/ACCESS.2020.3041738Muhammad, M., Armstrong, M., Elgendy, M.A.: A nonisolated interleaved boost converter for high‐voltage gain applications. IEEE J. Emerging Sel. Top. Power Electron. 4(2), 352–362 (2016). https://doi.org/10.1109/JESTPE.2015.2488839 http://www.deepdyve.com/assets/images/DeepDyve-Logo-lg.png "IET Generation, Transmission & Distribution" Wiley

Analysis, design and implementation of a high step‐up multi‐port non‐isolated converter with coupled inductor and soft switching for photovoltaic applications

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Abstract

INTRODUCTIONOne of the most important factors in the economic growth and development of a country in the modern world is the proper use and optimal consumption of energy [1–5]. In recent decades, the use of renewable energy has expanded significantly due to the declining and depletion of fossil fuels and concerns about environmental pollution [6, 7].One of the largest sources of energy for the earth is the sun, the energy of which is used in various ways [8, 9]. Solar power generation systems are mainly capable of generating energy during the day and incur high costs if energy storage is required for future use [10, 11]. Another type of system that feeds from renewable sources is wind turbines. One of the disadvantages of these systems is the need for installation in geographical windbreaks [12, 13]. Other energy generation systems include fuel cells. Fuel cells are a new technology for generating energy that produces high‐efficiency electrical energy from a direct combination of fuel and oxidizer without causing environmental and noise pollution. In fuel cells, hydrogen gas is mainly used as fuel and from its reaction with oxygen, in addition to electrical energy, water and heat are also produced. The main problems of fuel cells are the slow dynamics of the fuel cell and large changes in output voltage with load changes [14]. At the moment, due to the high cost of the main element of these systems, the use of this type of energy production system alone is not economical. Usually, it is necessary to use dc‐dc converters to adapt to variable output voltage, renewable energy sources with load voltage (dc bus) and also to manage power in these sources [15, 16]. By considering the strengths and weaknesses of each of these systems, it can be concluded that by combining energy production systems, in addition to using the strengths of each, their weaknesses can be minimized [17, 18]. Due to the variety of energy sources and the need to use more than one power source in some applications, it is better to use a multi‐input converter instead of several independent converters because in multi‐input converters use less semiconductor elements, less inductors and fewer capacitors. Therefore, its control structure is much simpler compared to several single‐input converters, which has increased the efficiency and increased the energy conversion capacity, as well as reduced the cost of manufacturing the converter [19, 20]. Due to the high importance of overall system efficiency in renewable energy production systems and also the limited space in many applications, increasing the efficiency and reducing the volume of these converters is very desirable. In many applications of multi‐input converters, especially in solar systems where the output voltage of the array terminal should not be more than a certain value, non‐isolated incremental converters can be used to provide high voltage gain and high efficiency at the same time [21].Types of multi‐input convertersIn a general classification, the types of multi‐input converters can be classified into two types of isolated multi‐input converters [22, 23]. and non‐isolated multi‐input converters [24, 25]. According to Figure 1. Isolation in multi‐input converters has advantages and disadvantages that should be selected based on the application of the system. Its advantage is the isolation of the sources from each other (in multi‐input magnetic converters) as well as the load of the sources, which allows the use of different sources with different voltages. Disadvantages of isolation include design problems for transformers with multiple windings as well as increased circuit volume. According to the above, the presence or absence of isolation depends entirely on the design of the system and is not merely a unique criterion for the suitability or inappropriateness of the converter [26]. In the non‐isolated converter topology, all ports have a common ground. High step‐up non‐isolation multi‐input dc‐dc converters are used in a variety of applications. Features of this converter are low number of parts, flexibility to increase the number of input sources, provide energy of power storage system and increase voltage gain. These converters use soft switching methods to increase the efficiency of the converter. One of the most important factors to consider when designing multi‐input converters is reducing the number of converter components. Non‐isolated multi‐input converters are more attractive for certain applications compared to isolated multi‐input converters due to the simplicity of structure and high efficiency as well as lower cost in applications where high voltage gain and isolation are not required [27].1FIGUREIsolation in multi‐port convertersNon‐isolated multi‐input converters are obtained by serial or parallel connection of base converters [28, 29]. Refs. [30] and [31] provide a complete analysis of how to extract different types of non‐isolated multi‐input converter structures. In this analysis, the types of non‐isolated multi‐input converter structures are classified into two general categories: pulse voltage source cell‐based (PVSC) multi‐input converters and pulse current cell‐based (PCSC) multi‐input converters. This analysis is the basis for extracting a variety of non‐isolated multi‐input converter structures.In ref. [32] different types of multi‐input converter topology with the help of these synthesis processes using PVSC and (PCSC) basic converters based on pulse width modulation (PWM) are shown. It has also been suggested that the current pulse source with the help of a boost converter can be suitable for high gain dc‐dc converters for hybrid electric vehicles.In ref. [33] a general method of synthesizing multi‐input converters using base converter cells, in other words PCSC and output filter cells is introduced. In this reference, a multi‐input converter with PCSC connection with a sample of output filter cells is proposed for which it does not require any energy buffer step. As a result, this topology saves on construction costs as well as simplifies design. Also, to better understand the above concepts, two design examples are given to illustrate the synthesis of multi‐input converters for use in power supply applications, Internet‐of‐Things (IOT) devices, and other applications.It can be said one of the most important factors that could be assumed in design and implementation of the most multi‐input converters is reducing the number of components.Soft switching in multi‐input convertersUsing multi‐input converters, it will be possible to remove some common elements in two separate converters. As a result, the number of general structure elements will decrease. So it will be relatively easier to provide soft switching in the resulting structure compared to two separate converters [34, 35].Based on the method used to provide soft switching in non‐isolated multi‐input converters, these converters classified into three general categories: non‐isolated resonant multi‐input converters [36], non‐isolated multi‐input converters with active clamp [37]. and non‐isolated multi‐input converters with soft switching [38].In multi‐input resonant converters with proper converter design, by adding additional elements, current and voltage pulses are transmitted in sinusoidal or quasi‐sinusoidal form. Using these methods, soft switching conditions and effective reduction of electromagnetic interference are provided. One of the problems with this method is the high stress current and voltage of the circuit elements. Also, in order to provide soft switching conditions for a wide range of load and input changes, there is mainly a need to use variable control methods, which will complicate the control circuit. In the multi‐input converter with active clamp, the at zero voltage switching conditions (ZVS) are provided by adding only one auxiliary switch and a small number of passive auxiliary elements [39]. In these converters, contrary resonant converters, the voltage stress and current stress of the elements are within the minimum value range. Also, it is possible to control the converter with conventional methods of controlling pulse width modulation and with constant frequency. However, the main problem with this method is the high eddy current of the converter. This eddy current is almost constant for a wide range of load changes, which reduces efficiency, especially at low powers. In multi‐input converters whose inputs are mainly renewable sources and also have large changes in production power during the day, this factor causes weakness in these converters.In soft‐switching multi‐input converters, zero‐voltage switching conditions are provided for a wide range of output power variations with minimal circulation current. Due to the advantages and disadvantages of the proposed methods, soft switching multi‐input converters have many advantages. Also, the voltage stress of the elements is minimal and the converter can be controlled by conventional on pulse width modulation control methods for a wide range of power and voltage changes. However, the main problem with multi‐input soft‐switching converters is the large number of auxiliary elements of the converter [40].Most energy sources are inherently low voltage and high step‐up structures are used to increase their voltage gain. To increase the voltage, gain, different methods are used in different converters according to their type and application. Some of these structures include coupled inductors, series capacitors in the power flow path, isolated transformers, and switched diode‐capacitor structures. Using these methods solves the problems associated with the high duty cycle in a conventional boost converter and improves the performance of the converter. Leakage inductance of coupled inductors is used to control the diode current drop and also to reduce the problem of reverse diode recovery and some soft switching methods are implemented by the coupled inductor to reduce switching losses. also the coupled inductor can act as a transformer to increase the voltage in non‐isolated high step‐up converters to avoid a very high duty cycle [41]. The capacitor is used as another voltage source to achieve a high gain [42].In general, it can be mentioned that the use of high frequencies in circuits increases switching losses and reduces circuit efficiency. Therefore, in order to reduce losses and increase the frequency, the use of soft circuit switching techniques is unavoidable. On the other hand, using soft switching techniques is a good solution to reduce RFI and EMI noise.Literature surveyHigh step‐up non‐isolated dc‐dc converters have a variety of applications. One of the most important features of these converters is the reduction of converter elements, flexibility to increase the number of input sources, power supply of power storage system and increase voltage gain [43, 44]. These converters use soft switching methods to increase the efficiency of the converter. One of the most important factors to consider when designing multi‐input converters is reducing the number of converter components.A general method for obtaining high step‐up non‐isolated dc‐dc converters using conventional dc‐dc converters with a circuit consisting of an inductor and voltage multiplier cells is presented in ref. [45].A high step‐up dc‐dc converter is proposed in ref. [46], Based on a combined structure of the coupled inductor and the switched‐capacitor, which simple structure and so low voltage switches help to increase the efficiency of the proposed converter. In addition, the voltage stress of capacitors and diodes is significantly reduced. Lack of soft switching and so the lack of multipolar capability of this topology are the problems of this proposed structure.A high‐gain dc‐dc converter with soft switching, based on the coupled inductor and switched‐capacitor technique, is provided in ref. [47]. capacitors are used in this converter that are charged in parallel and discharged in series with a coupled inductor to achieve high voltage gain. Using the energy stored in the inductance leakage of the coupled inductor, both switches act as ZVS, which increases efficiency. One of the disadvantages of this converter is the inability of the proposed multipolar topology.A high step‐up dc‐dc converter has been provided in ref. [48] that combines an inductor structure with a voltage doubling circuit to increase the voltage gain to achieve a high voltage gain. This proposed converter can be used effectively for the structure of a modular dc‐dc converter for solar cell systems. The only problem with this converter is the inability of the multipolar topology.A dc‐dc converter with high step‐up three‐ports is provided in ref. [49], which provides two separate paths for transferring power from the input source to the output load, in fact, for each input, it has a separate phase, so that the components work in each mode of operation changes. also It has a port for energy storage device such as batteries and ultra‐capacitors that each input source and energy storage device can provide a unique path of power current to the output load directly from the power source regardless of the load power status. Therefore, in order to reduce the number of converter components, some components are shared in different operating modes. the technique of coupled inductors is used to increase the voltage in this converter and also, to reduce the effect of leakage inductance and to create soft switching conditions, two active clamp circuits have been used. This type of converter has the ability to connect to other high step‐up converters, which include the coupled inductor and the active clamp structure. it turns them into multi‐input converters, which are achieved while providing soft switching mode and eliminating of leakage inductor effect.A non‐isolated high step‐up dc‐dc converter is proposed to combine the coupled inductor structure with the voltage multiplier and also the switched‐capacitor structure to increase the voltage. Soft switching methods are used to increase the converter efficiency, and switched capacitor cells and the proposed converter voltage multiplier circuits can reduce the voltage stress of switches and diodes. This converter is designed for clean energy applications and can be used well in high efficiency applications. One of its limited disadvantages is the high volume of the converter and the large number of converter elements, which complicates the converter and another disadvantage is the lack of multipolar capability of this topology [50].A non‐isolated high gain converter is presented in ref. [51]. which increases the voltage gain of the converter by using a coupled inductor and two switched‐capacitors that are charged in parallel by the inductor and discharged in series also reduces losses in the circuit. The only problem with this converter is the inability of the multipolar topology.Actually, the use of high step‐up techniques solves the problems associated with the high duty cycle in a conventional incremental converters and improves the performance of the converters.Main study and innovationIn this paper, the structure of a non‐isolated multi‐input converter has proposed, which can be increased by using a combination of coupled inductor technique with switched‐capacitor. To reduce the leakage inductance effects and provide soft switching mode, two active clamp circuits have been used in this converter. Simulation results in different conditions have presented to evaluate and analyse the performance of the proposed multi‐input converter. Also, a laboratory sample of this proposed converter has been made to evaluate the results, which shows the comparison of the correct design and analysis results. In summary, the innovation of the article can be expressed as follows:Using two active clamp circuits to reduce leakage inductance effects.Simple structure and increase the converter voltage gain.Using a combination of two methods, coupled inductor and switched‐capacitor.Using switches with low voltage stress resulting in low conductivity.Reduction of voltage stress in the proposed converter.Making a laboratory sample to review the analysis and design of the converter.Main structure of this paperThe structure of the article is as follows. After stating the importance of the issue in the introduction, describes the second part of proposed non‐isolated high step‐up dc‐dc converter and its eight operating modes. In the third part, the converter design is mentioned in four sections, which the computation of static voltage gains and the computation of voltage and current stress have expressed. In the fourth part, the results of converter simulation using ORACD software for different modes have shown. In the fifth part, the laboratory results of the proposed converter in EES discharge and charge modes have given. In the sixth part, the conclusion of the article is given.PROPOSED CONVERTER STRUCTURE AND CONVERTER OPERATING MODESIn multi‐input converters, the type of isolation is selected based on the type of system application, which its advantages include, isolation of sources from each other (in multi‐input magnetic converters). as well as the source load that provides the use of different sources with different voltages. In general, the problems of designing transformers with multiple windings as well as increasing the circuit volume are the disadvantages of isolation. Due to the high voltage gain of the proposed converter and also due to the high efficiency and simple structure in this type of applications, non‐isolated converters have more used. The main block diagram of the high step‐up multi‐input converter is shown in Figure 2. The non‐isolated high step‐up dc‐dc proposed converter structure is shown in Figure 3. According to the structure of the converter, it can be seen that in order to increase the voltage gain in the proposed converter, the combination of the structure of the coupled inductor with the voltage multiplier has been used. Also, to reduce the voltage stress of the main switch due to leakage inductor energy, active clamp circuits have been used. By combining these two methods, switches with low voltage stress and consequently low conductivity can be used. Adding a voltage multiplier cell increases the amount of leakage inductor and the transformer turns ratio, and as a result reducing the circuit volume. The performance of the proposed converter depends on the charging or discharging energy storage system (ESS) mode. The following is a complete analysis of each of the ESS charging or discharging modes.2FIGUREHigh step‐up multi‐input converter block diagram3FIGUREProposed non‐isolated high step‐up multi‐input converter structureESS discharging modeIn this case, diode D is always off and both phases operate independently of each other. Energy is transferring from input to output and in fact both of the phases operate completely independently of each other but similar to each other. S1 and S2 switches act as the main switch of converter. Two active clamp circuits, including the S3 switch and the CC1 clamp capacitor in the upper phase, and so S4 switch and the CC2 clamp capacitor in the lower phase, are used to reduce the voltage stress of the main switch due to the effects of recover the leakage inductance and also provide soft switching conditions.Also, elements C1, C2, L2, Dd1 and Dd3 in the upper phase and elements C3, C4, L4, Dd2 and Dd4 in the lower phase are used to increase the voltage gain. C1, C2, C3 and C4 are snubber capacitors and L1, L2, L3 and L4 are coupled inductors. Because in this case, both phases operation is completely similar to each other, only the upper‐phase operation modes have been investigated and analysed.But due to the exact similarity of the upper and lower phases in the ESS discharge mode, the operating modes for the lower phase are exactly the same as the upper phase. Figure 4 shows the main key waveform diagrams in the proposed converter. According to Figure 5, the proposed converter in one switching period has eight operating modes, which are expressed in the following equation of each mode:4FIGUREMain key waveform diagrams in the proposed converter in ESS discharging mode5FIGUREOperating modes in ESS discharging mode of proposed converterMode 1—Time range (t0 – t1)By the switch is turned on, the output diode current continues to flow. In this time range, equations are expressed as follows:1ilm(t0)=ilm(t0)+2Vc1−Vonlm(t−t0)\begin{equation}{i_{lm}}({t_0}) = {i_{lm}}({t_{\rm{0}}}) + \frac{{2\,{V_{c{\rm{1}}}} - {V_o}}}{{n\,{l_m}}}(t - {t_{\rm{0}}})\end{equation}2ilk(t0)+Vin−Vlmllk(t−t0)=ilk(t0)+Vin−2Vc1−Vonllk(t−t0)\begin{equation}{i_{lk}}({t_0}) + \frac{{{V_{in}} - {V_{lm}}}}{{{l_{lk}}}}(t - {t_{\rm{0}}}) = {i_{lk}}({t_0}) + \frac{{{V_{in}} - \left(\frac{{2\,{V_{c1}} - {V_{\rm{o}}}}}{n}\right)}}{{{l_{lk}}}}(t - {t_{\rm{0}}})\end{equation}where Vc1 and Vc2 are the multiplier capacitors (voltage amplifier), respectively. In general, C1 and C2 and L2 and diodes Dd3 and Dd1 form the voltage multiplier circuit together. The leakage inductor current is ilk and the magnetic induction current is ilm, and the voltages Vin, Vo, and Vlm are the input, output, and double‐ended voltages of the magnetic inductor, respectively. Also n indicates the turns ratio and LLK is leakage inductance.Mode 2—Time range (t1 – t2)This mode starts the moment the output diode turns off while the S1 is still on, the iLK current reaches iLm and the iLK exceeds iLm. In this case, the following equation is established:3ilm(t)=ilm(t1)+Vc1nlm(t−t1)\begin{equation}{i_{lm}}(t) = {i_{lm}}({t_1}) + \frac{{\,{V_{c{\rm{1}}}}}}{{n\,{l_m}}}(t - {t_1})\end{equation}4VLK=Vin−Vlm\begin{equation}{V_{LK}} = {V_{in}} - {V_{lm}}\end{equation}5iLK(t)=iLK(t1)+nVin−Vc1nLLK(t−t1)\begin{equation}{i_{LK}}(t) = {i_{LK}}({t_1}) + \frac{{n{V_{in}} - {V_{c1}}}}{{n\,{L_{LK}}}}(t - {t_1})\end{equation}Mode 3—Time range (t2–t3)When this mode starts, the main switch turns off under zero voltage switching (ZVS). Leakage inductance begins to resonance with capacitors Cs1 and Cs3. As a result, VCs1 increases and VCs3 decreases. For this mode, the equations are expressed as follows:6ilm(t)=ilm(t2)+vc1(nlm)(t−t2)\begin{equation}{i_{lm}}(t) = {i_{lm}}({t_2}) + \frac{{{v_{c1}}}}{{(n\,{l_m})}}(t - {t_2})\end{equation}7iLK(t)=1ZV1−vc1n+1ZCs3Cs1+Cs3(Vcs3(t2)−Vcc)×sin(ω(t−t2)+iLK(t2)cosω(t−t2\begin{eqnarray} {i_{LK}}(t) &=& \left[\dfrac{1}{Z}\left({V_1} - \dfrac{{{v_{c1}}}}{n}\right) + \dfrac{1}{Z}\dfrac{{{C_{s3}}}}{{{C_{s1}} + {C_{s3}}}}({V_{cs3}}({t_2}) - {V_{cc}})\right]\nonumber\\ &&\times\, \sin (\omega (t - {t_2}) + {i_{LK}}({t_2})\,\cos \omega (t - {t_2}\end{eqnarray}8Vcs1(t)=V1−vc1n+vc1n−V1+CS3CS1+CS3(Vcc−Vcs3(t2))×cos(ωt2)+[ilk(t2)Z]sinω(t−t2)\begin{eqnarray} {V_{cs1}}(t) &=& {V_1} - \dfrac{{{v_{c1}}}}{n}\nonumber\\ && +\, \left[\left(\dfrac{{{v_{c1}}}}{n} - {V_1}\right) + \dfrac{{{C_{S3}}}}{{{C_{S1}} + {C_{S3}}}}({V_{cc}} - {V_{cs3}}({t_2}))\right]\nonumber\\ &&\times\, \cos (\omega {t_2}) + [{i_{lk}}({t_2})Z]\,\sin \omega (t - {t_2})\end{eqnarray}Resonance impedance (Zo) and angular resonance frequency (ωo) are equal to:9Z0=LLK(Cs1+Cs3)\begin{equation}{Z_0} = \sqrt {\frac{{{L_{LK}}}}{{({C_{s1}} + {C_{s3}})}}} \end{equation}10ω0=1(Cs1+Cs2)LLK\begin{equation}{\omega _0} = \frac{1}{{\sqrt {({C_{s1}} + {C_{s2}})\,{L_{LK}}} }}\end{equation}Mode 4—Time range (t3–t4)This mode is a continuation of the previous mode and VCs3 reaches zero. VCs1 reaches VCc1 and from now on only the diode corresponding to the S3 switch works. at the end of this interval, iLK decreases and reaches iLm and decreases. The equations for this mode of operation are as follows:11ilm(t)=ilm(t3)+vc1nlm(t−t3)\begin{equation}{i_{lm}}(t) = {i_{lm}}({t_3}) + \frac{{{v_{c1}}}}{{n{l_m}}}(t - {t_3})\end{equation}12ilk(t)=vi−vc1n−VccLlk(t−t3)+ilk(t3)\begin{equation}{i_{lk}}(t) = \left(\frac{{{v_i} - \frac{{{v_{c1}}}}{n} - {V_{cc}}}}{{{L_{lk}}}}\right)(t - {t_3}) + {i_{lk}}({t_3})\end{equation}Mode 5—Time range (t4 – t5)Because the amount of iLK current is less than iLm in this mode, the direction of the current changes (the polarity of the coupled inductor changes). By changing the polarity, Dd1 and Dd3 are turned off and the output diode Do1 is biased directly. the equations are established in this mode below:13ilm(t)=−Vo+2Vc1+VccnLm(t−t4)+ilm(t4)\begin{equation}{i_{lm}}(t) = \frac{{ - {V_o} + 2\,{V_{c1}} + {V_{cc}}}}{{n\,{L_m}}}(t - {t_4}) + {i_{lm}}({t_4})\end{equation}14ilk(t)=−Vcc−−Vo+2Vc1+Vccn+V1Llk(t−t4)+ilk(t4)\begin{equation}{i_{lk}}(t) = \frac{{ - {V_{cc}} - \left(\frac{{ - {V_o} + 2\,{V_{c1}} + {V_{cc}}}}{n}\right) + \,{V_1}}}{{{L_{lk}}}}(t - {t_4})+ {i_{lk}}({t_4})\end{equation}Mode 6—Time range (t5 – t6)During this interval, while the body diode of S3 is operating, auxiliary switch S3 turns on under ZVS conditions. current will pass through the diode, as long as the switch current is positive and as soon as it becomes negative, current will pass through the switch. in this mode the equations are established below:15ilm(t)=−Vo+2vc1+vccnLm)(t−t5)+ilm(t5)\begin{equation}{i_{lm}}(t) = \frac{{ - {V_o} + 2\,{v_{c1}} + {v_{cc}}}}{{n\,{L_m}}})(t - {t_5}) + {i_{lm}}({t_5})\end{equation}16ilk(t)=−Vcc−−Vo+2vc1+vccn+V1Llk)(t−t5)+ilk(t5)\begin{equation}{i_{lk}}(t) = \frac{{ - {V_{cc}} - \left(\frac{{ - {V_o} + 2\,{v_{c1}} + {v_{cc}}}}{n}\right) + {V_1}}}{{{L_{lk}}}})(t - {t_5})+ {i_{lk}}({t_5})\end{equation}Mode 7—Time range (t6–t7)This mode starts when the auxiliary switch S3 is turned off under ZVS conditions. Capacitors Cs1 and Cs3 are discharged and the value of VCs3 increases and VCs1 decreases. the capacitor voltage of the main switch VCs1 reaches zero at the end of the end of this mode and the body diode begins to conduct. In this mode, equations are expressed as below:17ilm(t)=−Vo+2Vc1+VccnLm)(t−t6)+ilm(t6)\begin{equation}{i_{lm}}(t) = \frac{{ - {V_o} + 2\,{V_{c1}} + {V_{cc}}}}{{n\,{L_m}}})\,(t - {t_6}) + {i_{lm}}({t_6})\end{equation}18ilk(t)=−ilm(t6)n+1)+2Vc1−Vo+nV1Z(n+1)+Cs1Cs1+Cs3Vcs3(t6)Z+Cs3Cs1+Cs3VccZsinω(t−t6)+ilm(t6)n+1+ilk(t6)cosω(t−t6)\begin{eqnarray} {i_{lk}}(t) &=& - \dfrac{{{i_{lm}}({t_6})}}{{n + 1}}) + \left[\dfrac{{2\,{V_{c1}} - {V_o} + n\,{V_1}}}{{Z\,(n + 1)}}\,+ \dfrac{{{C_{s1}}}}{{{C_{s1}} + {C_{s3}}}}\dfrac{{{V_{cs3}}({t_6})}}{Z}\right.\nonumber\\ &&+\, \left.\dfrac{{{C_{s3}}}} {{{C_{s1}} + {C_{s3}}}} \dfrac{{{V_{cc}}}}{Z}\right]\,{\rm{sin}}\omega (t - {t_6})\nonumber\\ &&+\, \left[\frac{{{i_l}_m({t_6})}}{{n + 1}} + {i_{lk}}({t_6})\right]{\rm{cos}}\omega (t - {t_6}) \end{eqnarray}19Vc1(t)=Vo−2Vc1+nV1n+1+2Vc1−Vo+nV1n+1+Cs3Cs1+Cs3Vcc+Cs1Cs1+Cs3Vc3(t6)cosω(t−t6)+ilm(t6)n+1Z+ilk(t6)(n+1)ilm(t6)n+1sinω(t−t6)\begin{eqnarray} {V_{c1}}(t) &=& \dfrac{{{V_o} - 2\,{V_{c1}} + n\,{V_1}}}{{n + 1}} + \left[\dfrac{{2\,{V_{c1}} - {V_o} + n\,{V_1}}}{{n + 1}}+ \dfrac{{{C_{s3}}}}{{{C_{s1}} + {C_{s3}}}}{V_{cc}}\right.\nonumber\\ &&+\, \left.\dfrac{{{C_{s1}}}}{{{C_{s1}} + {C_{s3}}}}{V_{c3}}({t_6})\right]\,{\rm{cos}}\omega (t - {t_6}) + \left[\dfrac{{{i_{lm}}({t_6})}}{{n + 1}}Z\right.\nonumber\\ &&+\,\left. {i_{lk}}({t_6})\,(n + 1)\vphantom{\dfrac{{{i_{lm}}({t_6})}}{{n + 1}}}\right]\,{\rm{sin}}\omega (t - {t_6}) \end{eqnarray}In this mode, resonance impedance (Zo) and angular resonance frequency (ωo) are equal to:20Z0=LLK(Cs1+Cs3)\begin{equation}{Z_0} = \sqrt {\frac{{{L_{LK}}}}{{({C_{s1}} + {C_{s3}})}}} \end{equation}21ω0=n+1n(Cs1+Cs3)LLK\begin{equation}{\omega _0} = \frac{{n + 1}}{{n\,\sqrt {({C_{s1}} + {C_{s3}})\,{L_{LK}}} }}\end{equation}Mode 8—Time range (t7 – t8)The capacitor voltage of the main switch VCs1 reaches zero at the end of this mode and the body diode begins to conduct. The equations for this mode of operation are calculated as below:22ilm(t)=ilm(t7)+2Vc1−Vonlm(t−t7)\begin{equation}{i_{lm}}(t) = {i_{lm}}({t_7}) + \frac{{2\,{V_{c1}} - {V_o}}}{{n\,{l_m}}}(t - {t_7})\end{equation}23ilk(t)+Vin−Vlmllk(t−t0)=ilk(t7)+Vin−(2Vc1−Von)llk(t−t7)\begin{equation}{i_{lk}}(t) + \frac{{{V_{in}} - {V_{lm}}}}{{{l_{lk}}}}(t - {t_0}) = {i_{lk}}({t_7})+ \frac{{{V_{in}} - (\frac{{2\,{V_{c1}} - {V_o}}}{n})}}{{{l_{lk}}}}(t - {t_7})\end{equation}ESS charging modeIn ESS charging mode, the power generated by the upper source is much more than the required power of the load, and if ESS charging is required, additional energy is used to charge the ESS. In this mode, the proposed converter acts similarly to a buck converter and charging the ESS through the lower phase, and so the upper phase acting as the output load supplier. switching is done in the form of hard switching in this mode, because the converter performs as a buck converter and the buck converter by itself has an efficiency of over 90%. So hard switching is used, which does not complicate the circuit and has a proper efficiency in this mode. In this mode, S2 switch is always off and S4 acts as the main converter switch. Figure 6 shows the main waveform diagrams of the proposed converter in ESS charging mode. According to Figure 7, the proposed converter in ESS charging mode has 2 operating modes in one switching period.6FIGUREMain key waveform diagrams in the proposed converter in ESS charging mode7FIGUREOperating modes in ESS charging mode of proposed converterMode 1—Time range (t0 – t1)The time range in this mode is between t0 and t1. This mode starts where the S4 is still on. The current of iLm2 becomes positive and its value begins to increase. The equivalent circuit of this mode is shown in Figure 7a. In this case the following equations are established:24Vcc2=V1\begin{equation}{V_{cc2}} = {V_1}\end{equation}25VLm2=V1−V2\begin{equation}{V_{Lm2}} = {V_1}\; - {V_2}\end{equation}26iLm2t=iLm2t0+VLm2Lm2t−t0\begin{equation}{i_{Lm2}}\left( t \right)\; = \;{i_{Lm2}}\left( {{t_0}} \right) + \frac{{{V_{Lm2}}}}{{{L_{m2}}}}\left( {t - {t_0}} \right)\end{equation}27iLm2t=iLm2t0+V2−V1lm2t−t0\begin{equation}{i_{Lm2}}\left( t \right) = {i_{Lm2}}\left( {{t_0}} \right) + \frac{{{V_2} - {V_1}}}{{{l_{m2}}}}\left( {t - {t_0}} \right)\end{equation}Mode 2—Time range (t1 – t2)Time range in this mode is between t1 and t2. This mode starts where the S4 is on. Similar to the previous mode the current of iLm2 becomes positive and increase. The equivalent circuit of this mode is shown in Figure 7b. The equations for this operation mode are calculated as bellow:28VLm2=V2\begin{equation}{V_{Lm2}} = {V_2}\end{equation}29iLm2t=iLm2t1+VLm2Lm2t−t1\begin{equation}{i_{Lm2}}\left( t \right) = {i_{Lm2}}\left( {{t_1}} \right) + \frac{{{V_{Lm2}}}}{{{L_{m2}}}}\left( {t - {t_1}} \right)\end{equation}30iLm2t=iLm2t1+V2lm2t−t1\begin{equation}{i_{Lm2}}\left( t \right) = {i_{Lm2}}\left( {{t_1}} \right) + \frac{{{V_2}}}{{{l_{m2}}}}\left( {t - {t_1}} \right)\end{equation}HIGH STEP‐UP MULTI‐INPUT DC‐DC PROPOSED CONVERTER DESIGNIn ESS discharging modeThe design method along with the analysis of the proposed converter are presented in the following four sections. For the simplicity of the equations, all the elements of the circuit are ideally considered.Voltage gainWhen the main switch of each phase is on, the inductor Lm is charged by the input voltage Vin and the inductor voltage in the time interval from zero to DT is equal to:31Vlm=Vin\begin{equation}{V_{lm}} = {V_{in}}\end{equation}When the main switch is turned off, the inductor Lm starts to discharge, the voltage value of which is calculated in the time interval DT to 1‐DT as follows:32Vlm=(2n+1)Vin−Von+1\begin{equation}{V_{lm}} = \frac{{{\rm{(2}}n + {\rm{1)}}\,{V_{in}} - {V_o}}}{{n + {\rm{1}}}}\end{equation}Using the volt‐second balance, the voltage gain is given by:33VoVin=(2n+1)−nD1−D\begin{equation}\frac{{{V_o}}}{{{V_{in}}}} = \frac{{(2n + 1)\, - nD}}{{1 - D}}\end{equation}Voltage stressVoltage stress is calculated when the switch is off. Voltage stress of the main and auxiliary switches in the off state is equal to the voltage of the clamp capacitor.34Vsw3=Vsw1=Vcc=−nVin+Von+1\begin{equation}{V_{sw3}} = {V_{sw1}} = {V_{cc}} = - \frac{{n{V_{in}} + \,{V_o}}}{{n + 1}}\end{equation}The stress voltage of the output diode when the diode is off being calculated as bellow:35Vlm=nVin\begin{equation}{V_{lm}} = n{V_{in}}\end{equation}36VDo1=Vo−nVin\begin{equation}{V_{Do1}} = {V_o} - n{V_{in}}\end{equation}The voltage stress of diodes Dd3 and Dd1 is given by:37Vc1=Vc2=nVin\begin{equation}{V_{c1}} = {V_{c2}} = n{V_{in}}\end{equation}38Vlm=VL1\begin{equation}{V_{lm}} = V{L_1}\end{equation}39Vl2=nVlm\begin{equation}{V_{l2}} = n{V_{lm}}\end{equation}40VDd3=VDd1=−Vc1+V12=nVin−Vo\begin{equation}{V_{Dd3}} = {V_{Dd1}} = - \,{V_{c1}} + {V_{12}} = n\,{V_{in}} - {V_o}\end{equation}Current stressThe switch current stress is in fact the maximum current that passes through when the switch is off. It is calculated as follows:41Ismax=2(1+2n)−nDD(1−D)Io\begin{equation}{I_{s\max }} = 2\left[\frac{{(1 + 2\,n) - n\,D}}{{D\,(1 - D)}}\right]\,{I_o}\end{equation}The current stress of the output diode and diodes Dd3 and Dd1 are equal to:42IDomax=2Io1−D\begin{equation}{I_{Do\,max }} = \frac{{2\,{I_o}}}{{1 - D}}\end{equation}43IDd1max=IDd3max=2IoD\begin{equation}{I_{Dd1max}} = {I_{Dd3\,max }} = \frac{{2\,{I_o}}}{D}\end{equation}Design of the proposed converter elementsIn this section, the equations of the proposed converter elements, including output capacitors and coupled inductor, are described.Coupled inductor designIn most of the time, it is better to select the value of the inductor so that the converter has 20% of the current ripple on the inductor at the maximum amount of output power. Figure 8a shows the boundary conduction mode (BCM) and Figure 8b shows the inductor current waveform in continuous conduction mode (CCM). The minimum amount of magnetic inductance required to hold the converter in CCM mode can be expressed as bellow.44ΔIlm=20%Ilm\begin{equation}\Delta {I_{lm}} = 20\% \,{I_{lm}}\end{equation}45ΔIlm=2Ilm\begin{equation}\Delta {I_{lm}} = 2\,{I_{lm}}\end{equation}8FIGUREInductor current waveform. (a) In CCM mode and (b) in BCM modeMaximum load resistanceThe maximum load resistance (RlBCM) can be selected at about 20% of the nominal output power, in other words, it can be said to be five times the amount of output load. If p1 and p2 are upper phase power and lower phase power, respectively, the total power value (p) is equal to:46p=p1+p2\begin{equation}p = {p_1} + {p_2}\end{equation}47p=vo2RL\begin{equation}p = \frac{{v_o^2}}{{{R_L}}}\end{equation}48RlBCM=5RL\begin{equation}{R_{lBCM}} = 5\,{R_L}\end{equation}According to the nominal output power, the converter equivalent circuit and the average inductor current waveform in BCM mode are shown in Figures 9 and 10, respectively. The average inductor current is calculated as follows:49Ilm=Ilm(1−D)→Ilm=(n+1)Io(1−D)\begin{equation}{I_{lm}} = {I_{lm}}(1 - D) \to {I_{lm}} = \frac{{(n + 1)\,{I_o}}}{{(1 - D)}}\end{equation}50ΔIlm=vlmDTLm\begin{equation}\Delta {I_{lm}} = \frac{{{v_{lm}}D\,T}}{{{L_m}}}\end{equation}9FIGUREConverter equivalent circuit in BCM mode10FIGUREAverage inductor current waveform in BCM modeBy placing the above equations, the value of the inductor Lm is equal to:51Lm=D(1−D)2Rlbcm2(n+1)[(1+2n)−nD]fs\begin{equation}{L_m} = \frac{{D\,{{(1 - D)}^2}\,{R_{lbcm}}}}{{2\,(n + 1)\,[(1 + 2n) - nD]\,{f_s}}}\end{equation}Output capacitor designThe value of the output capacitor is basically selected according to the output voltage ripple. The legal output voltage ripple is considered to be 0.1% of the output voltage. due to the average current passing through the capacitor is zero, the output capacitor value is calculated according to the following equation:52Co≥Io·DΔVCo·fs\begin{equation}{C_o} \ge \frac{{{I_o}\cdot D}}{{\Delta {V_{Co}}\cdot {f_s}}}\end{equation}The value of the switch capacitor fundamentally depends on the voltage ripple and output power level. The values of capacitors C1 and C2 are determined from the following equation:53C1≥Io1−DΔVC1·fs\begin{equation}\;{C_1} \ge \frac{{{I_o}\left( {1 - D} \right)}}{{\Delta {V_{C1}}\cdot {f_s}}}\end{equation}The values of capacitors C1 and C2 are determined from the following equation:54Cc1≥VoΔVCc1·fs·RL\begin{equation}{C_{c1}} \ge \frac{{{V_o}}}{{\Delta {V_{{C_{c1}}}}\cdot {f_s}\cdot {R_L}}}\end{equation}Soft switchingIn power electronic converters, soft switching techniques are used to reduce losses by minimizing the overlap of the current and voltage of the switch during its on and off times. This paper uses the ZVS technique. Also the resonance phenomenon is used to zero the voltage across the two switches. It is clear that the turn‐off the switches conditions under ZVS are provided by snubber capacitors parallel to the switches. before switching, reverse current enters the body diode of the transistor and discharges the snubber capacitor as a result, the voltage across the two heads approaches zero, which provides the turn‐on conditions under ZVS. In this situation, the amount of energy stored in the leakage inductance must be more than the energy stored in the capacitor parallel to the switches:5512LLKiLK2(tx)≥12CsVds2(tx)\begin{equation}\frac{1}{2}{L_{LK}}i_{LK}^2({t_x}) \ge \,\frac{1}{2}{C_s}V_{ds}^2({t_x})\end{equation}5612LLK(n+1)Io1−D≥Cs1||Cs3vin1−D2\begin{equation}\frac{1}{2}{L_{LK}}\left[\frac{{(n + 1)\,{I_o}}}{{1 - D}}\right] \ge {C_{s1}}||\,{C_{s3}}\,{\left(\frac{{{v_{in}}}}{{1 - D}}\right)^2}\end{equation}According to the amount of load stored in the capacitors and their discharge current, the amount of dead time is calculated from the following equation.57Td>(c2+c4)V1ILmin\begin{equation}{T_d} > \,\frac{{({c_2} + {c_4})\,{V_1}}}{{{I_{L\min }}}}\end{equation}In ESS charging modeAs seen in the operating modes of the proposed converter in ESS charging mode, in this mode the converter works exactly the same as a buck converter, which includes elements of switches S2 and S4 and source V1 and diode D as well as inductor Lm2. The V2 storage source is charged by a buck converter. Given that in this operating mode the proposed converter performs similarly to a buck converter; the voltage gain as well as all the equations are exactly the same as the buck converter equations.SIMULATION RESULTSIn this section, the simulation results of the proposed converter using ORCOD software in different performance modes to confirm the theoretical analysis of the proposed converter are shown. The parameters of the proposed converter are given in Table 1. The simulation results are a function of the time shown and the time axis is in milliseconds.1TABLEParameter values in the proposed converterComponentsValuePower400 WFirst input voltage40 VSecond input voltage24 VOutput voltage400 VSwitching frequency100 kHzOutput capacitors38 μFSwitched capacitor4.7 μFClamp capacitor4.7 μFSwitchesIRFP260DiodesMUR840 and MUR460Simulation results in ESS discharge modeInput voltages are the main parameters for the design of the buck converter inductor to operate in BCM mode and provide soft switching conditions. Assuming the first input voltages (V1) and the second (V2) are 40 and 24 V, respectively, and the duty cycle (D) is greater than or equal to 0.5, and the value of the turns ratio for upper‐phase coupled inductors (n1) is 2.14. And for lower coupled inductor, (n2) is equal to 4.04. According to the output power and accurate calculation of RlBCM, the values of magnetic inductors in the upper (Lm1) and lower (Lm2) phases are equal to 150 and 57 μH, respectively. Leakage inductor and magnetic inductor current in the upper and lower phases are shown in Figures 11 and 12. As can be seen, the ripple magnetic inductor current is low, therefore the ohmic losses of the inductor and the switches are reduced, and as shown in the simulation results, the leakage inductance current at certain time intervals in the second section shows the negative values and when occurs the amount of iLK current becomes less than iLm and the direction of the current changes (the polarity of the mating inductor changes). Figure 13 and 14, show the current waveforms of the voltage multiplier capacitor and the clamp capacitor in the upper and lower phases, respectively. Figure 15, also shows the waveform of the output diode current in the upper and lower phases and shows the operating modes of the output diode correctly. As can be seen, at intervals when the auxiliary switch S3 is turned on under ZVS conditions, the current passes through the diode as long as the current of the switch is positive, and as soon as it is negative, the current will pass through the switch.11FIGURELeakage inductor current and magnetic inductor in upper phase12FIGURELeakage inductor current and magnetic inductor in lower phase13FIGUREVoltage switch‐capacitor current waveform in upper and lower phase14FIGUREClamp capacitor current waveform in upper and lower phase15FIGUREOutput diode current waveform in upper and lower phaseFigures 16 and 17 show the soft switching condition of the main and auxiliary switches of the upper and lower phases. As can be seen, the overlap of the current and voltage of the switch has reached its minimum value when it is turned on and off. Figure 18 shows the input current waveform from the 40 and 24 V source, respectively.16FIGURESoft switching condition of main keys (S1 and S2)17FIGURESoft switching condition of auxiliary keys (S3 and S4)18FIGUREInput current waveform 40 and 24 V sourcesSimulation results in ESS charging modeIn ESS charging mode, the input voltages are the main parameters for the design of the buck converter inductor to operate in BCM mode. Assuming the first input voltages (V1) and the second (V2) are 40 and 24 V, respectively, and the duty cycle (D) is greater than or equal to 0.5, and the value of the turns ratio for upper‐phase coupled inductors (n1) is 2.14. And for the lower phase coupled inductor (n2) is equal to 4.04. All values related to the design of the inductor and etc are similar to the discharge mode. The results of the simulation at full load are shown in Figures 19–27. Leakage inductance current in ESS charge mode is shown in Figure 19. gate‐source and drain‐source voltages on the switch S4 are given in Figures 20 and 21, respectively. The switch current waveform is shown in Figure 22. Gate‐source and drain‐source voltages on the switch S2 are shown in Figures 23 and 24, respectively. As can be seen, gate‐source voltage at switch S2 is zero, and in fact this switch is off. The switch S2 current waveform of can be seen in Figure 25. The low phase output diode waveform shown in Figure 26 display that no current is transmitted from the S2 switch to the output. Figure 27 shows the input current waveform of the source 40 and 24 V, respectively.19FIGUREInductor current waveform in ESS charging mode20FIGUREGate‐source voltage waveform of switch S4 in ESS charging mode21FIGUREDrain‐source voltage waveform of switch S4 in ESS charging mode22FIGURESwitch S4 current waveform in ESS charging mode23FIGUREGate‐source voltage waveform of switch S2 in ESS charging mode24FIGUREDrain‐source voltage waveform of switch S2 in ESS charging mode25FIGURESwitch S2 current waveform in ESS charging mode26FIGUREDiode output current waveform in lower phase in ESS charging mode27FIGUREInput current waveform of source 40 and 24 V in ESS charging modeA comparison between several similar converters with the proposed converter is given in Table 2. As can be seen, the proposed converter has a higher efficiency as well as higher output power compared to similar converters. As a result, in high step‐up applications, it can be used more than similar converters. In three converters, soft switching under zero voltage switching (ZVS) and in one converter, soft switching under zero current switching (ZCS) is used.2TABLEComparison of the proposed converter with similar convertersConverter topology[47][48][49][50][51]Proposed converterNumber of switch224224Number of Diode335627Voltage gainDNt+Ns+21−D$\frac{{D{N_t} + {N_s} + 2}}{{1 - D}}$1+2N−ND1−D$\frac{{1 + 2N - ND}}{{1 - D}}$1+N1−D$\frac{{1 + N}}{{1 - D}}$2+2N1−D$\frac{{2 + 2N}}{{1 - D}}$2+N1−D$\frac{{2 + N}}{{1 - D}}$(2N+1)−ND1−D$\frac{{( {2N + 1} ) - ND}}{{1 - D}}$Voltage stress on switchVs1−D$\frac{{{V_s}}}{{1 - D}}$Vo−NVin1+N$\frac{{{V_o} - N{V_{in}}}}{{1 + N}}$VoN+1$\frac{{{V_o}}}{{N + 1}}$Vo2(N+1)$\frac{{{V_o}}}{{2( {N + 1} )}}$Vo2+N$\frac{{{V_o}}}{{2 + N}}$−NVin+VoN+1$ - \frac{{N{V_{in}} + {V_o}}}{{N + 1}}$Voltage stress on output diodeDNt1−DVg$\frac{{D{N_t}}}{{1 - D}}{V_g}$(1+N)Vo1+2N−ND$\frac{{( {1 + N} ){V_o}}}{{1 + 2N - ND}}$Vo${V_o}$NVo(N+1)$\frac{{N{V_o}}}{{( {N + 1} )}}$(1+N)Vo2+N$\frac{{( {1 + N} ){V_o}}}{{2 + N}}$Vo−nVin${V_o} - n{V_{in}}$Multi‐input/port××√××√Soft switchingZVSZVSZVSZCSZVSZVSEnergy storage capability××√××√Method to reduce no. of componentsSoft switching techniquesActive clamp and combination of coupled inductor and switched capacitorIntegrating ESS charging and discharging paths with each otherCombines the advantages of switched capacitors, coupling inductors, and voltage multiplier techniquesZVS soft switching techniquesIntegrating ESS charging and discharging paths with each other and combination of the inductance coupling method with the voltage multiplierEfficiency96.29595.3394.794.897.63EXPERIMENTAL RESULTIn this section, the practical results obtained from the proposed converter in different operating modes are shown to confirm the theoretical analysis of the proposed converter. The laboratory sample structure of the proposed converter and how it is connected to the scope are shown in Figures 28 and 29.28FIGURELaboratory sample of the proposed converter29FIGUREConnecting the proposed converter to the load and scopeExperimental results in ESS discharging modeFigure 30 shows the soft switching condition in the main keys and Figure 31 shows the soft switching condition in the S3 and S4 keys in the ESS discharge mode. Figures 32 and 33 show the input current waveform of source 40 and 24 V, respectively.30FIGURESoft switching condition on S1 and S2 keys in ESS discharge mode31FIGURESoft switching condition on S1 and S2 keys in ESS discharge mode32FIGUREInput current waveform of source 4033FIGUREInput current waveform of source 24Experimental results in ESS charging modeFigure 34 shows the condition of the S2 switch in ESS charge mode. As can be seen, the S2 switch is off in this mode and only its diode conducts. Figure 35 shows the condition of the S4 key in ESS charge mode. Figure 36 and 37 show the input current waveform of source 40 and 24 V, respectively.34FIGURES2 key condition in ESS charging mode35FIGURES4 key condition in ESS charging mode36FIGUREInput current waveform of source 40 in ESS charging mode37FIGUREInput current waveform of source 40 in ESS charging modeAs the results obtained from the construction of the proposed converter show, the practical results completely confirm the simulation results. Figure 38 and 39 show the proposed converter efficiency under different output load conditions. As can be seen, in most load conditions, the proposed converter efficiency in the ESS discharge mode remains in the range of 97%. The maximum measured efficiency of this converter is 97.63%. In ESS charging mode, the proposed converter efficiency remains in the range of 95% and its maximum value is 95.37%. According to the results obtained from the simulation of the converter as well as the practical results obtained from the construction of the proposed converter, this converter has a suitable increase voltage gain and it has the highest possible efficiency in a range of loads and also all switches operate in all operating modes under soft switching conditions.38FIGUREThe efficiency of the proposed converter in different load conditions in ESS discharging mode39FIGUREThe efficiency of the proposed converter in different load conditions in ESS charging modeThe loss analysis of the proposed converter at full load is shown in Table 3 is given, which confirms the correctness of the proposed converter efficiency.3TABLEProposed converter analysisParametersLosses (MW)Do1543.37Do2818.24Dd1620.83Dd2645.69Dd3620.83Dd4645.69S12.92 × 103S25.8 × 103S3807.5S40.998 × 103According to the results obtained from the simulation of the proposed converter, some of the features and advantages of this converter can be described as follows:(a)Reduce the volume of the circuit.(b)Creating a soft switching condition for the active circuit elements, which has the ability to increase the switching frequency and reduce the volume of the converter.(c)Non‐isolated (same input and output ground, simple structure, no need for isolated transformer and low construction cost).(d)Efficiency increase of about 97.63%.CONCLUSIONIn this paper a high step‐up non‐isolated multi‐input converter is proposed, as we know, renewable energy sources are inherently low voltage, so high step‐up techniques are used to increase their voltage gain. Due to the non‐isolation of the proposed converter (the same input and output ground, no need for isolated transformers and also simple structure), the construction cost is lower compared to similar converters. Also, creating soft switching conditions for active circuit elements has increased the switching frequency and reduced the converter volume. This converter has the ability to connect to other high‐step‐up converters, which include the coupled inductor and the active clamp structure, and converts them into multi‐input converters. According to the simulation results, the proposed converter has an increase in the appropriate voltage gain and has the highest possible efficiency in a load range. Precisely the maximum measured efficiency of this converter is 97.63% and in most load conditions, the efficiency remains in the range of 97%, and also, all switches work in all operating modes under soft switching conditions. In order to confirm the results of the simulation of the proposed converter, a laboratory sample of the proposed converter was made. The practical results obtained from the proposed converter in different operating modes were presented to confirm the theoretical analysis of the proposed converter and the accuracy of construction was shown.FUNDING INFORMATIONThere are no funders to report for this submission.CONFLICT OF INTERESTThe authors declare that they have no known competing financial interests or personal relationships that could have appeared to influence the work reported in this paper.DATA AVAILABILITY STATEMENTData openly available in a public repository that issues datasets with DOIsREFERENCESFaiz, J., Shahgholian, G., Ehsan, M.: Stability analysis and simulation of a single‐phase voltage source UPS inverter with two‐stage cascade output filter. Eur. Trans. Electr. Power 18(1), 29–49 (2008). https://doi.org/10.1002/etep.160Borhani, M., Yaghoubi, S.: Improvement of energy dissipative particle dynamics method to increase accuracy. J. Therm. Anal. Calorim. 144, 2543–2555 (2021). https://doi.org/10.1007/s10973‐020‐10362‐1Shahgholian, G.: A brief review on microgrids: Operation, applications, modeling, and control. Int. Trans. Electr. Energy Syst. 31, e12885 (2021). https://doi.org/10.1002/2050‐7038.12885Niazvand, F., Kharrati, S., Khosravi, F., Rastgou, A.: Dual‐objectives energy and load management for an energy hub by considering diverse plannings and in the presence of CCUS technology and the TOU program. J. Intell. Proced. Electr. Technol. 14(54), 31–58 (2023). https://dorl.net/dor/20.1001.1.23223871.1402.14.54.3.9Shahgholian, G., Shafaghi, P.: State space modeling and eigenvalue analysis of the permanent magnet DC motor drive system. In: Proceeding of the IEEE/ICECT. Kuala Lumpur, Malaysia, pp. 63–67 (2010). https://doi.org/10.1109/ICECTECH.2010.5479987Kharrazi, A., Sreeram, V., Mishra, Y.: Assessment techniques of the impact of grid‐tied rooftop photovoltaic generation on the power quality of low voltage distribution network‐ A review. Renewable Sustainable Energy Rev. 120, 109643 (2020). https://doi.org/10.1016/j.rser.2019.109643Alam, M., Kumar, K., Dutta, V.: Implementation of multiple outputs fly back converter for dc microgrid application. In: Proceeding of the IEEE/ICESIP. Chennai, India, pp. 1–4 (2019). https://doi.org/10.1109/ICESIP46348.2019.8938288Jahanshahi Bavandpour, R., Ghadiri, H., Khodadadi, H.: Optimal design of a hybrid solar–wind–battery system using the grasshopper optimization algorithm for minimization of the loss of power supply probability. J. Intell. Proced. Electr. Technol. 13(51), 139–156 (2022). https://dorl.net/dor/20.1001.1.23223871.1401.13.51.9.2Yi, W., Ma, H., Peng, S., Liu, D., Ali, Z.M., Dampage, U., Hajjiah, A.: Analysis and implementation of multi‐port bidirectional converter for hybrid energy systems. Energy Rep. 8, 1538–1549 (2022). https://doi.org/10.1016/j.egyr.2021.12.068Xu, J., Liu, L., Wang, F.: Equilibrium strategy‐based economic‐reliable approach for day‐ahead scheduling towards solar‐wind‐gas hybrid power generation system: A case study from China. Energy. 240, 122728 (2022). https://doi.org/10.1016/j.energy.2021.122728Pal, S., Sinha, D.: Design and simulation of modified multi‐input interleaved boost converter for renewable resources. In: Proceeding of the IEEE/IEMRE. Kolkata, India, pp.1–6 (2021). https://doi.org/10.1109/IEMRE52042.2021.9386869Asy'ari, M.K., Musyafa, A.’, Indriawati, K.: Design of wind turbine output voltage control systems in multi‐input buck converter using fuzzy logic control for battery charging. In: Proceeding of the IEEE/ICAMIMIA). Batu, Malang, Indonesia, pp. 249–252 (2019). https://doi.org/10.1109/ICAMIMIA47173.2019.9223417Haghshenas, G., Mirtalaei, S.M.M., Mordmand, H., Shahgholian, G.: High step‐up boost‐flyback converter with soft switching for photovoltaic applications. J. Circuits, Syst. Comput. 28(1), 1–16 (2019). https://doi.org/10.1142/S0218126619500142Fernandez, A.M., Kandidayeni, M., Boulon, L., Chaoui, H.: An adaptive state machine based energy management strategy for a multi‐stack fuel cell hybrid electric vehicle. IEEE Trans. Veh. Technol. 69(1), 220–234 (2020). https://doi.org/10.1109/TVT.2019.2950558Güler, N., Irmak, E.: Design, implementation and model predictive based control of a mode‐changeable DC/DC converter for hybrid renewable energy systems. ISA Trans. 114, 485–498 (2021). https://doi.org/10.1016/j.isatra.2020.12.023Vidhya, S.D., Balaji, M.: Hybrid fuzzy PI controlled multi‐input DC/DC converter for electric vehicle application. Automatika 61(1), 79–91 (2020). https://doi.org/10.1080/00051144.2019.1684038Affam, A., Buswig, Y.M., Othman, A.K.B.H., Julai, N.B., Qays, O.: A review of multiple input DC‐DC converter topologies linked with hybrid electric vehicles and renewable energy systems. Renewable Sustainable Energy Rev. 135, 110186 (2021). https://doi.org/10.1016/j.rser.2020.110186Sharifiyana, O., Dehghani, M., Shahgholian, G., Mirtalaei, S.M.M., Jabbari, M.: An overview of the structure and improvement of the main parameters of non‐isolated dc/dc boost converters. J. Intell. Proced. Electr. Technol. 12(48), 1–29 (2022). https://dorl.net/dor/20.1001.1.23223871.1400.12.48.6.6Lee, S., Kang, B.: Quasi‐resonant passive snubber for improving power convers ion efficiency of a dc–dc step‐down converter. IEEE Trans. Power Electron. 33, 2026–2034 (2018). https://doi.org/10.1109/TPEL.2017.2698211Rajan, P., Jeevananthan, S.: A new partially isolated hybrid output of multiport multilevel converter for photovoltaic based power supplies. J. Energy Storage 45, 103436 (2022). https://doi.org/10.1016/j.est.2021.103436Revathi, B.S., Mahalingam, P., Gonzalez‐Longatt, F.: Interleaved high gain DC‐DC converter for integrating solar PV source to DC bus. Sol. Energy. 188, 924–934 (2019). https://doi.org/10.1016/j.solener.2019.06.072Reddi, N.K., Ramteke, M.R., Suryawanshi, H.M., Kothapalli, K., Gawande, S.P.: An isolated multi‐input ZCS DC–DC front‐end‐converter based multilevel inverter for the integration of renewable energy sources. IEEE Trans. Ind. Appl. 54(1), 494–504 (2018). https://doi.org/10.1109/TIA.2017.2753160Nawaz, A., Wu, J., Long, C.: Mitigation of circulating currents for proportional current sharing and voltage stability of isolated DC microgrid. Electr. Power Syst. Res. 180, 106123 (2020). https://doi.org/10.1016/j.epsr.2019.106123Rani, P.H., Navasree, S., George, S., Ashok, S.: Fuzzy logic supervisory controller for multi‐input non‐isolated DC to DC converter connected to DC grid. Int. J. Electr. Power Energy Syst. 112, 49–60 (2019). https://doi.org/10.1016/j.ijepes.2019.04.018Amaleswari, R., Prabhakar, M.: Non‐isolated multi‐input dc‐dc converter with current sharing mechanism. Int. J. Electron. 108(2), 237–263 (2021). https://doi.org/10.1080/00207217.2020.1789760Reddy, K.J., Natarajan, S.: Energy sources and multi‐input dc‐dc converters used in hybrid electric vehicle applications– A review. Int. J. Hydrogen Energy. 43(36), 17387–17408 (2018). https://doi.org/10.1016/j.ijhydene.2018.07.076Zhang, N., Sutanto, D., Muttaqi, K.M.: A review of topologies of three‐port dc–dc converters for the integration of renewable energy and energy storage system. Renewable Sustainable Energy Rev. 56), 388–401 (2016). https://doi.org/10.1016/j.rser.2015.11.079Vidhya, S.D., Balaji, M.: Hybrid fuzzy PI controlled multi‐input DC/DC converter for electric vehicle application. Automatika. 61(1), 79–91 (2020). https://doi.org/10.1080/00051144.2019.1684038Affam, A., Buswig, Y.M., Othman, A.B.H., Julai, N.B., Qays, O.: A review of multiple input DC‐DC converter topologies linked with hybrid electric vehicles and renewable energy systems. Renewable Sustainable Energy Rev. 135, 110186 (2021). https://doi.org/10.1016/j.rser.2020.110186Liu, Y., Chen, Y.: A systematic approach to synthesizing multi‐input DC–DC converters. IEEE Trans. Power Electron. 24(1), 116–127 (2009). https://doi.org/10.1109/TPEL.2008.2009170Bairabathina, S., Balamurugan, S.: Review on non‐isolated multi‐input step‐up converters for grid‐independent hybrid electric vehicles. Int. J. Hydrogen Energy. 45(41), 21687–21713 (2020). https://doi.org/10.1016/j.ijhydene.2020.05.277Chen, G., Liu, Y., Qing, X., Wang, F.: Synthesis of integrated multiport dc–dc converters with reduced switches. IEEE Trans. Ind. Electron. 67(6), 4536–4546 (2020). https://doi.org/10.1109/TIE.2019.2931214Shan, Z., Ding, X., Jatskevich, J., Tse, C.K.: Synthesis of multi‐input multi‐output dc/dc converters without energy buffer stages. IEEE Trans. Circuits Syst. II: Express Briefs. 68(2), 712–716 (2021). https://doi.org/10.1109/TIE.2019.2931214Mohammadi, M.R., Farzanehfard, H.: Family of soft‐switching bidirectional converters with extended ZVS range. IEEE Trans. Ind. Electron. 64(9), 7000–7008 (2017). https://doi.org/10.1109/TIE.2017.2686308Elserougi, A., Abdelsalam, I., Massoud, A., Ahmed, S.: A bidirectional non‐isolated hybrid modular DC–DC converter with zero‐voltage switching. Electr. Power Syst. Res. 167, 277–289 (2019). https://doi.org/10.1016/J.EPSR.2018.11.009Kanathipan, K., Moury, S., Lam, J.: A fast and accurate maximum power point tracker for a multi‐input converter with wide range of soft‐switching operation for solar energy systems. In: Proceeding of the IEEE/APEC. Tampa, FL, pp. 2076–2083 (2017). https://doi.org/10.1109/APEC.2017.7930985Kwon, J.M., Kwon, B.H.: High step‐up active‐clamp converter with input‐current doubler and output‐voltage doubler for fuel cell power systems. IEEE Trans. Power Electron. 24(1), 108–115 (2009). https://doi.org/10.1109/TPEL.2008.2006268Faraji, R., Ding, L., Rahimi, T., Kheshti, M., Islam, M.R.: Soft‐switched three‐port dc‐dc converter with simple auxiliary circuit. IEEE Access. 9, 66738–66750 (2021). https://doi.org/10.1109/ACCESS.2021.3076183Salehi, N., Mirtalaei, S.M.M., Mirenayat, S.H.: A high step‐up dc–dc soft‐switched converter using coupled inductor and switched capacitor. Int. J. Electron. Lett. 6(3), 260–271 (2018). https://doi.org/10.1080/21681724.2017.1357195Reddi, N.K., Ramteke, M.R., Suryawanshi, H.M., Kothapalli, K., Gawande, S.P.: An isolated multi‐input ZCS DC–DC front‐end‐converter based multilevel inverter for the integration of renewable energy sources. IEEE Trans. Ind. Appl. 54(1), 494–504 (2018). https://doi.org/10.1109/TIA.2017.2753160Li, W., He, X.: Review of nonisolated high‐step‐Up DC/DC converters in photovoltaic grid‐connected applications. IEEE Trans. Ind. Electron. 58(4), 1239–1250 (2011). https://doi.org/10.1109/TIE.2010.2049715Andrade, J.M., Coelho, R.F., Lazzarin, T.B.: High step‐up dc–dc converter based on modified active switched‐inductor and switched‐capacitor cells. IET Power Electron. 13(14), 3127–3137 (2020). https://doi.org/10.1049/iet‐pel.2020.0064Waseem, M., Saeed, L., Khan, M.Y.A., Saleem, J., Majid, A.: A multi input multi output bidirectional DC‐DC boost converter with backup battery port. In: Proceeding of the IEEE/ICPESG. Mirpur Azad Kashmir, Pakistan, pp. 1–6 (2018). https://doi.org/10.1109/ICPESG.2018.8384526Revathi, B.S., Mahalingam, P., Gonzalez‐Longatt, F.: Interleaved high gain dc‐dc converter for integrating solar PV source to dc bus. Sol. Energy 188, 924–934 (2019). https://doi.org/10.1016/j.solener.2019.06.072Schmitz, L., Martins, D.C., Coelho, R.F.: Comprehensive conception of high step‐up dc–dc converters with coupled inductor and voltage multipliers techniques. IEEE Trans. Circuits Syst. I: Regul. Pap. 67(6), 2140–2151 (2020). https://doi.org/10.1109/TCSI.2020.2973154Ye, Y., Chen, S., Yi, Y.: Switched‐capacitor and coupled‐inductor‐based high step‐up converter with improved voltage gain. IEEE J. Emerging Sel. Top. Power Electron. 9(1), 754–764 (2021). https://doi.org/10.1109/JESTPE.2020.2971525Zhang, X., et al.: Novel high step‐up soft‐switching dc–dc converter based on switched capacitor and coupled inductor. IEEE Trans. Power Electron. 35(9), 9471–9481 (2020). https://doi.org/10.1109/TPEL.2020.2972583Sathyan, S., Suryawanshi, H.M., Ballal, M.S., Shitole, A.B.: Soft‐switching dc–dc converter for distributed energy sources with high step‐up voltage capability. IEEE Trans. Ind. Electron. 62(11), 7039–7050 (2015). https://doi.org/10.1109/TIE.2015.2448515Faraji, R., Farzanehfard, H.: Soft‐switched nonisolated high step‐up three‐port dc–dc converter for hybrid energy systems. IEEE Trans. Power Electron. 33(12), 10101–10111 (2018). https://doi.org/10.1109/TPEL.2018.2791840Seo, S.W., Ryu, J.H., Kim, Y., Choi, H.H.: Non‐isolated high step‐up dc/dc converter with coupled inductor and switched capacitor. IEEE Access. 8, 217108–217122 (2020). https://doi.org/10.1109/ACCESS.2020.3041738Muhammad, M., Armstrong, M., Elgendy, M.A.: A nonisolated interleaved boost converter for high‐voltage gain applications. IEEE J. Emerging Sel. Top. Power Electron. 4(2), 352–362 (2016). https://doi.org/10.1109/JESTPE.2015.2488839

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